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V LIBRARY
2 Michigan State
23 .‘O UI liverSity

 

 

 

This is to certify that the
thesis entitled

INVESTIGATION OF TRACTION DRIVE SYSTEMS FOR
SERIES HYBRID ELECTRIC BUSES

presented by
Craig B. Rogers

has been accepted towards fulfillment
of the requirements for the

MS. degree in Electrical Engineering

 

 

fiwpfls"

Major mm Signature

Date

MSU is an Affirmative Action/Equal Opportunity Employer

PLACE IN RETURN BOX to remove this checkout from your record.
To AVOID FINES return on or before date due.
MAY BE RECALLED with earlier due date if requested.

 

DATE DUE DATE DUE DATE DUE

 

 

 

 

 

 

 

 

 

 

 

 

 

 

 

5/08 Kthrolecc8-PrelelRC/DateDueJndd

 

INVESTIGATION OF TRACTION DRIVE SYSTEMS FOR SERIES HYBRID
ELECTRIC BUSES

By

Craig B. Rogers

A THESIS

Submitted to
Michigan State University
in partial fulfillment of the requirements
for the degree of

MASTER OF SCIENCE

Electrical Engineering

2009

ABSTRACT

INVESTIGATION OF TRACTION DRIVE SYSTEMS FOR SERIES HYBRID

ELECTRIC BUSES.
By

Craig B. Rogers

Abstract—This paper presents a comparison of power electronics topology designs for
implementation in a Series Hybrid Electric Bus project. This specific hybrid vehicle
design uses an internal combustion diesel engine (ICE), an ac electric generator, a
passive/active rectifier/inverter, power storage, a traction motor inverter, and a traction
motor/brake. The value of the hybrid vehicle comes from its increased efficiencies. The
hybrid system decouples the ICE’s power and speed from that of the bus. This
decoupling is accomplished by replacing the traditional mechanical or hydro mechanical
transmission with power electronics and energy storage facilities. This decoupling allows
ICE operation at its most efficient power versus speed point. During normal operation,
the energy storage system can be used to handle the dynamic loads and the internal
combustion engine can handle the steady loading. This hybrid system also recaptures
braking energy which is lost with conventional bus systems. The power electronics
facilitate the process via control of power flow, battery management, motor control and
generator control. There are two presently existing inverter/converter configurations that
can be designed to implement these power electronics solutions. There is also a new
solution using the novel Z-Source inverter as a current source inverter. Each of these
designs have specific operation modes, control methods and features that are analyzed
and discussed in this paper. The advantages and disadvantages of each are discussed.

The concepts are then demonstrated via simulation and experimental results.

DEDICATION
Dedicated to my Brother
John Russell Rogers

June 09, 1961 - February 24, 2008

iii

ACKNOWLEDGEMENTS
I would like to thank my parents for their guidance love and support throughout the years.
Without them and their instilled values of perseverance, this work would not have been

possible.

I would further like to thank my advisor Dr. Peng for his support, guidance and
encouragement. Thank you for all that you have taught me in the years that I have known

you.

I would also like to thank Dr. Strangas for the guidance and education that he has

bestowed upon me.

Thank you to my children Nicole, Angela, and Brian for their love, understanding, and

being a part of the best parts of my life.

And finally, thank you to my colleagues at Michigan State University’s Power

Electronics and Motor Drives Laboratory. I have greatly enjoyed your friendship and

appreciate your shared knowledge.

iv

TABLE OF CONTENTS

 

 

 

 

 

 

 

 

 

 

 

 

 

 

 

 

 

 

 

 

 

 

 

LIST OF FIGURES _ - vii
1. INTRODUCTION - - . -- . - - -_ - -- - - . -- 1
1.2 Conventional Vehicles ...... - - .......... 1
1.3 Alternative Vehicles ..... -- 4
1.4 Fuel Cell Vehicles 14
1.5 Outline of Thesis -- - l6
2. HYBRID ELECTRIC VEHICLE Power Electronics Review 18
2.1 Voltage Source Inverter - 18
2.1.1 Switching Analysis of the Voltage Source Inverter _ 21
2.1.2 Voltage Source Inverter Control - 25

2.2 Current Source Inverter 32
2.3 Z-source Inverter 34
2.3.2 Maximum Boost Control - - 44
2.3.3 Maximum Constant Boost Control 48

2.4 Input/Output Equations of the Z-source Inverter 55

3. INTRODUCTION TO POWER ELECTRONICS T OPOLOGIES AND
THEIR OPERATING PRINCIPLES - 57
3.1. Traditional Input Rectifier/V SI to Output VSI - 68
3.1.1 Battery & Traction Motor Mode 68
3.1.2 Battery, Starter Motor, Traction Motor Mode 69
3.1.3 Generator, Battery Discharge, Traction Motor Mode - 71
3.1.4 Generator, Battery Charging, Traction Motor Mode - 72
3.1.5 Regenerative Braking and Battery Charging Mode 72
3.1.6 High voltage battery - 75
3.1.7 Low voltage battery with DC/DC converter 75

 

3.1.8 Comparison of the high voltage battery vs. the low voltage battery.. 77

3.2 Input Rectifier/VS] to Output CSI with Z-Source impedance network

(battery in parallel with Z’s capacitor)- - . -- -- - - -- 82

 

 

 

 

 

 

 

 

 

 

 

 

 

 

 

 

 

 

3.2.1 High voltage battery .................. - - _ - 85
3.2.2 Low voltage battery with DC/DC converter - - 86
3.3 Input Rectifier/C81 to Output VSI with Z-Source impedance network
(battery in parallel with Z’s capacitor) - 86
3.3.1 High voltage & Low voltage battery options. 89
3.4 Input Rectifier/V SI to Output CSI with Z-Source impedance network
(battery in parallel with Input Converter) _ 89
3.4.1 High voltage- - 93
3.4.2 Low voltage battery with DC/DC converter 93
3.5 Summary- - ...... -- - . ................ 94
COMPARISON OF POWER ELECTRONICS TOPOLOGIES .................. 95

4.2 Description of Switching Device Power Comparison - 95
4.3 DC/DC Boost — Buck Converter 97
4.4 Traditional VSI with PWM -- 100
4.5 Traditional Z-Source Inverter [ the Voltage Fed Z-Source Inverter (VF-
ZSI) ] with PWM - 102
4.6 Current Fed Z-Source Inverter (CF-ZSI). 105
4.7 Application of Equations to the Back to Back Topologies _ - 117
4.7.1 Back to Back VSI-to—VSI Network Topology -- 117
4.7.2 Back to Back VSI-to-CSI Z—Network Topology (V SI-Z-CSI) .......... 119
4.7.3 Back to Back CSI-to—VSI Z-Network Topology - - 125
4.7.4 Comparison of Circuit Topologies 129
4.8 Future Work - 138

 

vi

LIST OF FIGURES

 

 

 

 

 

 

 

 

 

 

 

 

 

 

 

 

Figure 1 Typical Internal Combustion Engine (ICE) Vehicle- - 3
Figure 2 Typical Electric Vehicle (EV) - - - 5
Figure 3 Typical Series HEV [5] - -- - - 9
Figure 4 Typical Parallel HEV [5] - - ................... - ...... - 12
Figure 5 Typical Series-Parallel HEV [6] - - - _ - - 13
Figure 6 Typical Series-Parallel HEV with Planetary Gear Set [6] -- 14
Figure 7 Typical Fuel Cell Vehicle [7] - -- ....... - 15
Figure 8 (A) IGBT Module Example [7] (B) IGBT Module Example [7] ................. 19
Figure 9 Traditional voltage source inverter 20
Figure 10 Single phase half-bridge inverter - - -- - - -- 21
Figure 11 Single Phase Inverter Voltage Output - _ 22
Figure 12 Single Phase Inverter Voltage Output — loomed In - -- - _ 23
Figure 13 Single Phase Inverter Voltage Output at Zero Crossing - - - 24
Figure 14 Typical Three Phase Voltage Source Inverter -- -- -- - 25
Figure 15 Simple pulse width modulator [12] ..... 26
Figure 16 SPWM generation - - 27
Figure 17 Saturated Pulse Width Modulation- - 28
Figure 18 Standard sine wave reference signal at 100% modulation index [14] ...... 30

Figure 19 Reference signal modified with third harmonics, equivalent to standard
sine wave reference signal at 100% modulation index [14] -- - - -_ 31

 

vii

Figure 20 Reference signal modified with third harmonics at 100% modulation

 

 

 

 

 

 

 

 

 

 

 

 

 

 

index [14] - -- -- - -_ - _ -- -- 31
Figure 21 Traditional current source inverter.. ...... -- -- 32
Figure 22 Current Source Inverter with Gate Turn Off Thyristors 33
Figure 23 Z-source inverter -_ ....... -- - ..... - ......... 34
Figure 24 Illustration of the 100 switching vector - -- - 36
Figure 25 Illustration of the 011 switching vector -- 36
Figure 26 Illustration of the V“. switching vector ..... 37
Figure 27 Traditional PWM Control Method - -- - - _ 38
Figure 28 281 During SCZ State ..... -- -- _ 40
Figure 29 Modified PWM with shoot through zero states [9] -- - -- 40
Figure 30 Voltage gain of the simple boost control [16] ............. 42
Figure 31 Z-Source Inverter in Open Circuit Zero Mode - - 43
Figure 32 Maximum boost control strategy [16] - -- - -- -- 45
Figure 33 Voltage gain of the maximum boost control strategy [16] _ 46
Figure 34 Maximum boost control with third harmonic injection strategy [16] ...... 47

Figure 35 Voltage gain of the maximum boost control with third harmonic injection

 

 

 

I16] -- -- - 48
Figure 36 Maximum constant boost control strategy [17] -- - 49
Figure 37 Voltage gain of the maximum constant boost control strategy [17] ......... 51
Figure 38 Third Harmonic Injection into Reference Signals -- - - -- 52
Figure 39 Third Harmonic Injection with Max. Constant Boost PWM [17] ............ 53

Figure 40 Voltage gain of the maximum constant boost control with third harmonic
injection [17] _ 54

 

viii

Figure 41 Voltage stress comparison of different control methods [17] .................... 55
Figure 42 Traditional Input Rectifier/V SI to Output VSI .......................................... 61

Figure 43 Input Rectifier/VSI to Output CSI with Z—Source impedance network
(battery in parallel with Z’s capacitor)--- - - _- - - - -- 63

Figure 44 Input Rectifier/CSI to Output VSI with Z—Source impedance network
(battery in parallel with Z’s capacitor) ------ - 65

Figure 45 Input Rectifier/VSI to Output CSI with Z-Source impedance network
(battery in parallel with Input Converter) - 67

 

Figure 46 A single phase model of the Voltage Source Inverter and the Generator 70

 

 

 

 

 

 

 

 

Figure 47 Voltage Phase Angle Control---- - - - - -- -- . 73
Figure 48 Voltage Phase Angle Control--- - - 74
Figure 49 DC/DC Converter ------------------------------------------ - - 76
Figure 50 Series Combination of Battery Cells;See Note - - 79
Figure 51 Back to Back VSI, CSI with Z-Source Network..-"- - 82
Figure 52 Back to Back VSI topology ----------------- - - 83
Figure 53 Back to Back VSI, CSI Z-Source Topology ................... - 84
Figure 54 VSI-Z-CSI Regenerative Mode with VSI in SCZ 85
Figure 55 CSI-Z-VSI Topology ..................................................................................... 86
Figure 56 Shoot-Through Duty Cycle vs. DC Voltage Across Inverter ..................... 89
Figure 57 Input VSI/Rectifier to Output CSI topology - - 90

 

Figure 58 Regenerative Braking with VSI-Z-CSI and Battery Parallel to VSI Active
State 91

 

Figure 59 Regenerative Braking with VSI-Z-CSI and Battery Parallel to VSI Open
Circuit Zero State - - - 92

Figure 60 Regenerative Braking with VSI-Z-CSI and Battery Parallel to VSI, Short
Circuit Zero State - 92

 

ix

Figure 61 Input VSI to Output CSI with Z-Source DC Link ..................................... 93

 

 

Figure 62 DC/DC Boost-Buck Converter ..................................................................... 98
Figure 63 Traditional VSI ............................................................................................ 100
Figure 64 Traditional Z-Source Inverter .................................................................... 102
Figure 65 Current Fed ZSI .......................................................................................... 105
Figure 66 IGBT Module Schematic ............................................................................ 106
Figure 67 Reverse Blocking IGBT Module Schematic (RB-IGBT) -- - -- 106
Figure 68 RB-IGBT Symbol ........................................................................................ 106
Figure 70 SCZ Switch States of VF-ZSI ..................................................................... 109
Figure 71 SCZ Regions of the CF-ZSI ........................................................................ 111
Figure 72 SCZ Switch States of CF-ZSI detail....................... -- - .... - - - 115
Figure 73 Back to Back VSI-to—VSI Network Topology ........................................... 117
Figure 74 VSI-Z-CSI .................................................................................................... 120
Figure 75 VSI-Z-CSI; CSI in OCZ State .................................................................... 121
Figure 76 VSI-Z—CSI; CSI in SCZ State ..................................................................... 121
Figure 77 CSI-Z—VSI .................................................................................................... 125
Figure 78 ILZ Average ................................................................................................. 131
Figure 79 SDP Comparison ......................................................................................... 135
Figure 80 SDP (Normalized) Comparison ................................................... 121
Figure 81 Proposed Topology .............................................................. 134
Figure 82 Possible Implementation of the Proposed Topology (1) .................... 134
Figure 83 Possible Implementation of the Proposed Topology (2) .................... 135
Figure 84 Possible Implementation of the Proposed Topology (3) .................... 135

KEYS TO SYMBOLS OR ABBREVIATIONS

ac voltage at the traction motor
'Peak ( I}TM )
-RMS ( V m )

-Line to Line (VTM-LL)
ac voltage at the generator/ starter motor
-peak (VG)
-RMS ( VG)
-Line to Line (VOLL)
Average Line Current (1 AV or Iav)

Battery terminal voltage (VBAT)

Boost factor (B)
Buck—boost factor ( B B )
Capacitor voltage (VC)

Constant power speed ratio (CPSR)
Current Fed Z-Source Inverter (CF-ZSI)

Current Source Inverter (CSI)

dc-link voltage across the generator inverter bridge (Vd) or (V905)

dc-link voltage across the traction motor inverter bridge (Voom)

Electric vehicle (EV)

xi

Electro Magnetic Interference (EMI)
Fuel cell (F C)
Fuel cell hybrid electric vehicle (F C HEV)

Fuel cell vehicle (FCV)

Generator Power, Three Phase ( P3415 )

Hybrid electric vehicle (HEV)

Hybrid Vehicle (HV)

Hybrid vehicle (HV)

Insulated Gate Bi-Polar Junction Transistor Module (IGBT)

Internal combustion engine (ICE)

Lower shoot-through envelope (V n)

Lower shoot-through envelope reference signal (Vm-ef)

Modulation index (M)

Open Circuit Zero Switching Period (Tocz) or (T1)

Peak Line Current (i LINE)
Phase A Lower switch gate control signal (San)
Phase A upper switch gate control signal (Sap)

Phase A voltage reference (Va‘)

Phase angle between phase voltage and line current (8)

Phase B Lower switch gate control signal (3an

xii

Phase B upper switch gate control signal (pr)

Phase B voltage reference (Vb*)

Phase C Lower switch gate control signal (Sen)
Phase C upper switch gate control signal (Sop)

Phase C voltage reference (VJ)

Pulse width modulation (PWM)

Reverse Blocking Insulated Gate Bi-Polar Junction Transistor Module (RB-IGBT)
Series Hybrid Electric Vehicle (SHEV)

Shoot-through duty cycle (D0)

Short Circuit Zero Switching Period (Tscz)

Sinusoidal pulse width modulation (SPWM)

State of charge (SOC)

Switching cycle interval (T)

Switching Device Power (SDP)

Time during shoot through (To)
Traction Motor Power, Three Phase ( P3¢_TM )
Upper shoot-through envelope (VP)

Upper shoot-through envelope reference signal (Vpref)

Voltage Fed Z-Source Inverter (VF-ZSI)
Voltage gain (MB)

Voltage Source Inverter (VSI)

xiii

Zero emission vehicle (ZEV)
Z-Source Inductor Current (ILz)

Z-Source Inverter (ZSI)

xiv

1. INTRODUCTION

Alternatively powered vehicles are becoming of great importance. Several factors are
causing the need fro change from traditional vehicles. As the population of the planet
continues to grow, and as the industrialization of these populations continue to grow, the
rate of depletion of traditional fuel sources (fossil fuels — oil, and coal primarily) is also
increasing. Along with this increased global demand for power comes the increased
pollution from consuming fuels. Both the decreasing supply of fossil fuels and the
increasing level of pollutants in our environment are becoming very significant concerns.
As a result, there is a growing search for ways in which these problems may be reduced
or solved. Both alternative fuels and ways to increase efficiency in the use of fuels are
being sought after. One such approach is the development of hybrid vehicles which are
much more fuel efficient and which produce much less pollutants. This introduction will
provide a discussion of traditional vehicle, electric vehicles, fuel cell vehicles, hybrid
vehicles. The hybrid vehicle discussion is broken down into parallel, series,

series/parallel, and finally a discussion of the series hybrid electric bus is presented.

1.2 Conventional Vehicles
The tradition of fuel power vehicles started in 1769 with a self propelled military tractor
invented by the French engineer and mechanic, Nicolas Joseph Cugnot (1725-1804).
Much before this date, conceptual designs had been made by both Leonardo Da Vinci
and Isaac Newton. [1] In 1823, Samuel Brown patented the first internal combustion

engine to be applied industrially. It was compression-less and based on the "Leonardo

cycle," In 1838, a patent was granted to William Bamet (English) for an engine using
in-cylinder compression. In 1854, the Italians Eugenio Barsanti and Felice Matteucci
patented the first working efficient internal combustion engine in London but did not go
into production with it. It was similar in concept to the successful Otto Langen indirect
engine, but not so well worked out in detail. [2]. Various fuels were used over these
years. By far the most prevalent designs today use gasoline or diesel fuel. Present day
buses are primarily users of diesel fire]. The typical road vehicle in use today is based
upon a standard platform. That is a gasoline or diesel filCICd internal combustion
compression engine. The engine is coupled to the remainder of the drive train through a
multiple gear ratio transmission. From the transmission, power is delivered to the drive
wheels through a drive shaft/s and a differential gear box. The engines are of the
cylindrical compression type, and the transmission usually has three to five discrete gear
ratios. The function of the differential gear box is to allow one of the drive wheels to
rotate at a different speed than the other drive wheel while both are delivering power to
the road. Each of these drive train components is mechanical. See Ffigure 1 for a

schematic of the typical conventional vehicle.

 

 

 

J Mechanical
Power Flow

—>

 

 

 

 

l
Transmission
l

 

 

 

 

Differential
Gear Box

   

 

Figure 1 Typical Internal Combustion Engine (ICE) Vehicle

Hybrid vehicles use some combination of these mechanical components as well as
electrical components to drive the vehicle. Hybrid busses, due to their use in urban areas
and due to their particular drive cycles, are excellent candidates for use of hybrid

technology.

1.3 Alternative Vehicles
Various countries and governments across the globe are trying g to foster the
development of alternative vehicles that mitigate some of these concerns. Alternatives
are necessary to eventually create vehicles that do not further deplete crude oil reserves
and that produce pollutants and a much reduced rate. The population of the world
continues to grow, more nations are becoming industrialized, and burning of fossil fuels
is believed to contribute to global warming. There are many reasons to turn to alternative
that would alleviate some or all of these concerns. One such alternative is the electric
vehicle (EV). This vehicle does not have an internal combustio9n engine and does not
burn any fuels on board. Its power is usually received from the electric utility and stored

in batteries that are on board the vehicle.

 

Battery —}

 

 

 

 

 

Mechanical
40 Power Flow
1 FL
I Electric

l
l Controller Power Flow

l

 

 

!X
i I
l l
K7

Motor

 

 

 

 

Figure 2 Typical Electric Vehicle (EV)

The stored power is then applied to eclectic motors to drive the vehicle rather than using
an ICE. This type of vehicle is very efficient as compared to conventional vehicles.
Presently, the fuel cost per mile (cost per kilo-watt hour (kWhr)) is about 3 cents per mile
as compared to 9 cents per mile for gasoline in the conventional vehicle. One of the main
advantages of the EV is that it is a zero emission vehicle (ZEV). Although there is some
pollution created when the electricity is generated at the power plant, this is insignificant

compared to emissions from conventional ICE vehicles. EVs are also very efficient when

compared to conventional ICE vehicles; only about 20% of the chemical energy in
gasoline is converted into work at the wheels of an [C E vehicle, 75% or more of the
energy from a battery reaches the wheels of an EV [3]. EVs also provide strong
performance, due to the fact that electric motors provide nearly peak power even at low
speeds. Cost, performance, and convenience of an EV all depend on the battery. There
are several types of batteries commonly used in EVs, advanced lead-acid batteries,
nickel-metal hydride, and lithium polymer batteries [3].The draw back to this type of
vehicle is that it takes more time to recharge the batteries and recharging stations are not
yet commonly available. Because of the limited storage capacity of the batteries, the

range of these vehicles is also more limited.

1.3 Hybrid Vehicles

Hybrid vehicles (HV) are growing in popularity for their fuel efficiency and lower
emissions. Fuel costs are increasing and the world’s fuel reserves are continuing to grow
smaller. Since the world’s supply of crude oil is based in just a few countries, a nation’s
dependence on oil also makes it dependent upon the oil producing nation. Also, pollution
issues are increasing in weight. These are the main reasons why the demand for HVs is
increasing and why governments are encouraging the sales and development of these

vehicles.

A HV is any vehicle that combines two or more sources of power to propel the

vehicle. Some of the first HVs were busses, trains, and submarines. Hybrid methods are

now being applied to passenger cars and small trucks as well. There are three main
configurations of HVs. They are Series Hybrid, Parallel Hybrid, and Series/Parallel

Hybrid. They are described and discussed below.

A common type of hybrid electric vehicle (HEV) combines the energy storage of the EV
with an energy source such as an ICE. HEVs have higher efficiency levels due to
regenerative braking, and due to their ability to control power from the ICE. By
controlling the power flow, it is possible to utilize the most efficient points of operation.
Regenerative braking takes advantage of the concept that energy can neither be created
nor destroyed — it just changes form. The conventional vehicle converts the kinetic
energy of the moving vehicle into heat energy when the brakes are applied to slow the
vehicle. This radiated heat is lost to the atmosphere. The HEV converts this inertia into
electrical energy by using the traction motor as an electric generator. The production of
electrical energy by the generator also produces a resistive torque against the free rotation
of the drive wheels which slows the vehicle. This electrical energy is stored as potential
energy in the vehicle’s battery bank and or ultra capacitors. In this way, energy that
would have been wasted is saved for later use. Also, the HEV can use a smaller ICE.
The ICE only needs to supply the consistent steady state load while the battery and
electric motor can provide power for the dynamic load. Some examples of the dynamic
load would be the additional power needed to accelerate and to climb hills. The smaller
ICE uses less fuel that the larger engines do. The smaller engine requires less fuel into
the smaller cylinders to achieve the correct air to fuel ratios. Also, an ICE has moving
parts which involve friction. The smaller parts have less surface fraction. Also, the

engines have linear reciprocating motion components which must undergo four

accelerations for each crank shaft revolution. Obviously, the lower mass components

require less energy.

1.3.1 Series Hybrid Vehicle

The series HEV delivers all of the power to the drive wheels via electric power. None of
the power is delivered with direct mechanical connections. For example, a typical
configuration uses an ICE connected to a generator. The generator is connected to the
traction motor through an electric power inverter and conductors. The power inverter is
usually connected to power storage devices such as batteries and or ultra capacitors. See

Figure 3.

 

 

ICE Battery ’

 

 

 

 

 

 

 

 

 

Mechanical

(X Power Flow

:7 Electric

R Power Flow

Generator 1 2 Controller

if
V

Electric
Motor

 

 

 

 

 

 

 

 

 

 

 

Gear

 

 

 

 

 

Figure 3 Typical Series HEV [4]

The series HEV uses an ICE which runs for extended periods of time at
maximum efficiency, and shuts off when the batteries reach full charge. The ICE is then
allowed to operate at the average required power level and the batteries will accept power
when this average exceeds instantaneous power demand. The battery likewise provides
power when the power required exceeds the average power provided.

In the series HEV configuration the electric motors and power electronics must be

sized for the whole load whereas the parallel case can use a smaller inverter and motor.
The ICE is smaller that the conventional vehicle and is comparable to that of the parallel
case. Also, this design does not require the additional equipment needed for the parallel
case. In higher power applications (busses, freight trains, subway trains), this is a very
effective design application.

The series HEV suffers due to the number of energy conversions. The engine
converts petroleum potential energy to kinetic energy (rotational motion); the generator
converts this kinetic energy to electrical energy. The electrical energy is then converted
to chemical energy in the battery and to mechanical (kinetic energy) in the traction motor.
However, the series HEV does not require a lossy hydro mechanical transmission. Also,
batteries have limited life expectancy. New battery technology has made great
improvements with this and there are several batteries that have long life periods when
the battery state of charge is maintained with in a fairly narrow band and when the charge

and discharge rates are maintained within specified limits [3].

_ 1.3.2 Parallel Hybrid Vehicle

The parallel hybrid vehicle has two power flow paths. One is mechanical from the ICE
to the wheels, and the other is electrical from the generator and batteries to the electric
motor and then mechanically to the wheels. The battery is recharged from regenerative
braking and by operating the ICE at a power level in excess of the instantaneous power
demand. This excess power is delivered to the batteries through the transmission, to the
motor (being operated as a generator), to the power inverter/converter to the battery. The

main advantages of this system are that both the engine and the motor size can be

10

minimized since both can deliver power at the same time to accommodate the peak load
requirements. Also, there are potentially less energy conversions for the mechanical
power flow path. On the other hand this system typically utilizes a hydro mechanical

transmission which has losses, and the control is more complicated than the series HEV.

[3].

11

 

Mechanical

 

 

 

 

 

 

 

 

 

 

 

 

 

 

 

Power Flow
Battery

Electric

Power Flow 4
[CE Controller
, , Electric
Transmrssron
Motor

 

 

 

 

 

 

 

 

 

Gear Wheel

   

 

 

 

Figure 4 Typical Parallel HEV [4]

12

1.3.3 Series-Parallel Hybrid Vehicle

The series-parallel HEV combines features from both of the preceding cases. This
system can Operate in series mechanical mode, series electrical mode, or any combined
ratio of the two. As a result, the control system can choose the most advantageous
operation mode for the given circumstances. Figure 5. shows a power flow diagram for a
series-parallel hybrid. Also notice the planetary component in this figure. This is also a
very significant part of the efficiency of the design. The planetary gears (see figure 6. for
a diagram) deliver power from the ICE to the wheels or to the generator. It also couples
power transfer to or from the generator. The generator is an instrumental part of this
control. The generator is used to create either positive or negative torque so that the
desired gear ratio is achieved. As a result of the ideal gear ratio, the ICE is continuously
operated at its most efficient point for the given power demand. Also, the planetary gear

system does not use lossy hydraulic torque converters as the conventional vehicle does.

p
to wheels

 

 

 

fuel + tea) - ~ 1.0),.

—.I engine

ut= 150$

   

counter

 

 

 

 

Figure 5 Typical Series-Parallel HEV [5]

13

   
     

2
A

”drive shtdrive_sh

A1 ‘\
5

N1 drive shaft

l““‘1 '1';

Figure 6 Typical Series-Parallel HEV with Planetary Gear Set [5]

This system benefits from the reduced size ICE, generator, power inverter, and traction
motor due to its parallel firnctionality. Also, the operating system is able to select the

most advantageous operation mode for the given circumstances

1.4 Fuel Cell Vehicles

Fuel cell vehicles (FC HEV) are still being developed and have not yet achieved
commercial production. However, this type of vehicle offers a great deal of benefits in
terms of efficiency, and very low emissions. Its environmental impact is similar to that

of an EV, yet the FC HEV has a large range comparable to the conventional EV. The FC

14

HEV utilizes fuels such as hydrogen, methanol, or gasoline to produce electrical power
without combustion. The waist product of the reaction is typically H20 and C02. The
drive train of a FC HEV thus becomes: Fuel Cell, Inverter, and Traction Motor. Most
designs will likely include a battery as well. The typical fuel cell does not have the
ability to respond quickly to power demand changes. Therefore, the fuel cell can be used
to supply average power and the battery can be used to supply or absorb the difference
between average power and instantaneous power. Adding regenerative braking to the F C
HEV makes the design very fuel efficient. Also, the infrastructure for these fuels does

not yet exist. Below is a diagram of the FC HEV structure.

 

! 1
Fuel Cell

|
j Mechanical

 

 

 

 

. I Power Flow
‘ I
<7
Electric

l
l

Battery KL—i> Controller PowerFIOW
j .fl

 

 

 

 

L/_I/j< F
l

J E
\-

Motor

 

   

 

 

 

Figure 7 Typical Fuel Cell Vehicle [6]

15

1.5 Outline of Thesis

The goal of this research is to investigate the feasible options available for the
design of traction drive systems for series hybrid electric buses. The traction drive system
will deliver power to and from the ICE via a generator and it will deliver power to and
from the traction motor. This system will also include interface and control of storage
batteries and or ultra capacitors. The following topologies will be considered:

0 Back to back voltage source inverters/converters

Current source inverter (C SI) to Voltage source inverter (VSI) coupled with the Z-

Source network -- OR --

o VSI to CSI coupled with the Z-Source network

0 High voltage batteries or low voltage batteries with a bi-directional DC-DC buck
boost converter

0 Each configuration will be analyzed (operating principle, design, and control).

0 A comparison of these systems will be included.

Chapter 2: Reviews the current HEV power electronics technologies, and introduces
their control strategies. The Z-source inverter and its control strategies are also
introduced.
Chapter 3: Introduction to Power Electronics Topologies and their operating principles
[delivery of power from the ICE and generator to the traction motor]
3.1. Traditional Input Rectifier/V SI to Output VSI
i. High voltage battery

ii. Low voltage battery with DC/DC converter

16

3.2.

ii.

3.3.

ii.

3.4.

ii.

Input Rectifier/VSI to Output C SI with Z-Source impedance network
(battery in parallel with Z’s capacitor)
High voltage battery

Low voltage battery with DC/DC converter

Input Rectifier/C SI to Output VSI with Z-Source impedance network
(battery in parallel with Z’s capacitor)
High voltage battery

Low voltage battery withy DC/DC converter

Input Rectifier/V SI to Output CSI with Z-Source impedance network
(battery in parallel with Input Converter)
High voltage

Low voltage battery with DC/DC converter

Chapter 4. Comparison of Power Electronics Topologies, Switch Device Power Ratio,

Design and Control of the Power Electronics Topologies, Strengths and Weaknesses,

Conclusions Related to the Topologies, Future Work

17

2. HYBRID ELECTRIC VEHICLE Power
Electronics Review

HEVs use an inverter to convert DC voltage and current to AC voltage and
current. The AC voltage applied to the traction motor can be varied in frequency to
change the motor speed, varied in magnitude to deliver power smoothly at low medium
and high power settings, and varied in phase angle to control power flow to the traction
motor or from the motor when it is being used as a brake. The traditional method for
making this conversion is to use a voltage source inverter or a current source inverter.
Another method that is still somewhat new and which offers great features is the use of a

Z-source inverter. Each of these topologies will be discussed.

2.1Voltage Source Inverter

The two traditional topologies are the voltage source inverter and the current
source inverter. The voltage source is shown in figure 9. the basic structure uses a dc
voltage source connected in parallel with a large capacitor. The capacitor makes up the
dc link that connects the voltage source to the three phase bridge inverter. The inverter
uses 6 insulated gate bi-polar transistors which include an anti-parallel diode (IGBT

module)(fig. 8).

18

External __c_9_r_r_r_r_ection

Aux C Vic:
IE1

Aux E

 

External connection

 

4H]
$1

 

e I] '
¢ll¢fl a?
E5)
67

 

 

 

 

 

 

 

 

 

 

 

 

 

Outline type code: F

 

 

 

(B)
Figure 8 (A) IGBT Module Example [7] (B) IGBT Module Example [7]

l9

A characteristic of the voltage source inverter is a smooth voltage supply and a current

supply with ripple.[7]. Below is a schematic of a three phase voltage source inverter.

Three Phase Inverter

 

DC

Voltage DC ——
Source UNI——
Capacitor

 

 

 

 

 

\ \ \To Traction Motor

 

 

 

 

 

Figure 9 Traditional voltage source inverter

The voltage source inverter has the following limitations: The ac output voltage
must be less than the dc link voltage. The voltage source topology can only buck voltage
levels. Also, this inverter is not allowed to have a top switch and a bottom switch on
simultaneously in the same phase leg. Such an occurrence will cause destructively high
short circuit currents through the switching devices. This potential result leaves the
inverter vulnerable to electromagnetic interference. Adding a DC-DC converter to the
design can resolve the first issue [8]. However, this adds additional cost and complexity
to the circuit as well as a decrease in efficiency. The second limitations can be resolved
by adding dead time to the switching scheme to create a safety margin between the time
that one switch in a phase leg turns off and the next one turns on. However, the addition

of dead time adds harmonic distortion to the output wave forms.

20

2.1.1 Switching Analysis of the Voltage Source Inverter
The three phase voltage source inverter is similar to the single phase version

shown in figure 10. The three phase version includes two more switch sets in parallel [9]
with the one (T1 and T2) shown below and the switching functions of the other parallel
phase legs are phase shified by 1200 from the other. First the single phase version will be

discussed.

 

 

 

VS/ 2

 

 

VS/2

 

T2)

 

 

 

Figure 10 Single phase half-bridge inverter

When the top switch (T1) is gated on, the load has Vs/2 impressed across it, while
the switch T2 is being gated on impresses a voltage of —Vs/2 across the load. By
controlling the switches, the average output voltage can be controlled to approximate a
sinusoidal wave form. The following three graphs show the output wave forms “Vout”
and the sinusoidal approximation “Vout —— average”. The first graph shows a whole cycle

and the second two graphs are zoomed in at a couple of different points so that the details

21

of the wave form can be seen.

 

Figure 11 Single Phase Inverter Voltage Output

22

323a

 

a:

0.806 wmood omood mmood

 
 
 

-----;.----a

38.0

0.8m-

0.0

0.03

odom

Odom

Figure 12 Single Phase Inverter Voltage Output — Zoomed In

23

 

Figure 13 Single Phase Inverter Voltage Output at Zero Crossing

24

 

(Sf

2.1.2 Voltage Source Inverter Control

Below is a schematic of a typical three phase voltage source inverter.

\\\
I \/\J

\\\

I l J

 

 

U'D)
P

 

O

 

 

 

 

 

 

Figure 14 Typical Three Phase Voltage Source Inverter

There are many different pulse width modulation (PWM) methods available to control
the three phase inverter. These control methods are algorithms designed for controlling
the state of the six switches in the inverter. The traditional V-source inverter has six
active vectors (or switching states) when the dc voltage is impressed across the load and
two zero vectors when the load terminals are shorted through either the upper three
switching devices or the lower three switching devices. The total of eight switching states
(six active and two zero) are used in the traditional pulse width modulation (PWM)
control methods. One common example of sinusoidal PWM (SPWM) uses sinusoidal

reference voltages compared to a triangle carrier wave. See figure 15 below.

The pulse width modulator produces a logic signal, referred to as sinusoidal PWM

(SPWM) that is used to command the inverters power transistors to turn on (saturated)

25

and off (cutoff), in such a way as to produce a sine wave at the inverter’s output. As
figure 15 shows a simple pulse width modulator can be implemented using a comparator.
The comparator is used to compare a reference signal, Vref, and a carrier signal, Vcarrier.
The comparator produces a logic-level output as shown in figure 16, which is logic-high
whenever Vrcf is greater than Vcam-cr, and logic-low whenever Vref is less than Vcarrier [10].
This output is used to control the upper switch in a phase leg. To control the lower switch

the signal is inverted.

 

 

 

 

 

SPWM
\I ll lllll H II II
Switch
- gate
Comparator signal
Vref Vcarrier
’\/ WV Triangle
Wave
:3 ? 10kHz

Figure 15 Simple pulse width modulator [10]

26

v_...\ A
» VW it
.

Switch gate signal
\-** _ I] ll fl

Figure 16 SPWM generation

 

 

 

 

 

 

 

 

 

 

 

 

 

 

 

This comparator output is then connected to the gate driver circuit of the IGBT modules.
The logic high output therefore is used to turn on (close the switch) the IGBT and the low
logic signal is used to turn it off. By proper control of the V_ref signal; frequency, phase

angle, and amplitude of the inverter’s output can be controlled.

The amplitude of the inverter’s output can be controlled by controlling the amplitude of
the V_ref signal. As the V_ref signal is increased in magnitude, V_ref will be greater
than V_carrier for longer periods of time and the IGBT switch will be closed for longer
periods of time. As a result, the average magnitude of the invert’s output is higher. The
ratio of V_ref to V_carrier is called the modulation index [9]. When V_refs peak value
is equal to the peak value of V_carrier, this is called a modulation index of 1 or 100%.
At this modulation level, the inverter is outputting the highest amplitude sinusoidal
voltage that it can with this particular control method. If the modulation index is
increased above 100%, the comparator output would remain at the high logic state for a
greater period of time, but the linear relationship between V_ref and the average output

signal will be lost [11], as shown in figure 17.

27

/\ /l/\/\ /\ A
W/ \/ \/ \/ \/ VVW

Switch gate si nal
i n l

v "I
Figure 17 Saturated Pulse Width Modulation

 

 

 

 

 

 

 

 

The maximum available voltage on a motor using a standard three-phase power inverter
is approximately 86% of the dc link voltage [12]. Figure 18 shows the standard sine wave
reference signal, which can achieve 100% (of V_carrier) maximum modulation index.
Third harmonic injection is a method used to increase the modulation index above 100%.
With this method, a sinusoidal voltage at three times the frequency of V_ref is added to
the V_ref signal. The resulting reference signal has shorter rise and fall times and a
flattened peak. See figure 19. Since the peak is flattened, we can now increase the
magnitude of V_ref above 100% of V_carrier. This will allow the logic high state time
period to be increased which will increase the inverter’s average output voltage level. In

the example shown in figure 19, the coefficient of 1/6 is chosen for the third harmonic

signal.
V _ref = II? _reflsin(a),,_,eft + (0)

=0

A

3rd

A

_. ref

 

 

 

 

28

 

sin(3a)V_ ”11+ go)

I(

V3rr! : (’éjil} _ ref

The combined signal is:

.3 am

With the chosen coefficient of 1/6, the fundamental amplitude (V

sin(a)V_reft + (0)

 

Sin(3a)V_reft + (P) + it; ref

 

) can be

_ ref

 

increased by 15% [13]. This method is particularly appealing for 3 phase system
applications. As can be seen from figure 19, the new combined reference voltage is no
longer completely sinusoidal and therefore, the inverter output of a single phase is
likewise changed. However, the wave shape effect of the third harmonic injection is
eliminated in the line to line voltages of a balanced three phase inverter. This is because
a 1200 shift of the fundamental is a 3600 shift of the third harmonic. The net result is

simply that the inverter is now able to output a higher level of its dc link voltage.

29

 

 

 

 

 

 

 

A Sinewave Modulation 3
es: s
'2 80 a:
E l’ \ >
o l a
a: 40 3
2 20 \ / g
3

'g 0 V 5
E I

 

 

 

Figure 18 Standard sine wave reference signal at 100% modulation index [12]

3O

3rd Harmonic Injection

 

 

 

 

 

 

 

x100 a: o
E 80 m 33 'E _
t: 7? . - o (B
- 60%W \ E E ’E
S ’ X y"\. KT“ '6 iii 3
"' 40 * .4“ 0 I
3 20 '5 .'= 1:
13 \W/ g .c: :I
E 0 -- I- u.
_ M. -____-_-| i I

 

 

Figure 19 Reference signal modified with third harmonics, equivalent to

standard sine wave reference signal at 100% modulation index [12]

3rd Harmonic Injection

 

‘k

100 ‘3

 

 

 

 

 

 

 

 

Modulation Index (%)

— Modified Reference
+ Third Harmonic
-- Fundamental

 

 

 

Figure 20 Reference signal modified with third harmonics at 100% modulation
index [12]

31

2.2 Current Source Inverter

Figure 21. below shows the traditional current source inverter (CSI).

: w

_I _J _I

 

 

 

reverse blocking _I
IGBT (RB- IGBT)

: To Load

 

Figure 21 Traditional current source inverter

One exception may be that the reverse blocking IGBT (RB-IGBT) is a new
device. In the past, the typical switching device of the current source inverter was either

the thyristor, or the Gate Turn Off Thyristor in series with a power diode.

32

 

To Load

 

 

 

 

 

3232p

GTO
Thyristor

Figure 22 Current Source Inverter with Gate Turn Off Thyristors

In more recent years a RB-IGBT module has been used with the current source inverter.
The main features of the CSI are the current source and the one directional current flow
in the inverter. The current source is usually a voltage source in series with a large
inductor. Since inductors cannot change their current magnitude instantaneously, the
current source inverter provides a steady level of current. Current on the dc side is
smoothed by an inductance, while the voltage on the dc side contains ac ripple [7]. A
main advantage of the current source inverter is the smooth flow of current that they
provide. A disadvantage, however, is that this type of inverter cannot withstand an open
circuit condition in its inverter [9]. Such an un-allowed condition could exist when all
top switches or all bottom switches remain open even for a very short period of time. In
the event of an open circuit condition (which could result from miss-gating due to EMI,
etc.) the current source will push current through the switching devices even though they
are in their high resistance off state. This current produces destructively high
temperatures in the devices. A safe guard against this condition is provided by providing

overlap is the switching scheme. Each switch is caused to remain closed for an extended

33

period of time that overlaps with the time that the next device is turned on. This method
is usually effective although it does cause some additional harmonics in the waveforms.
Another problem that the current source inverter has is that it can only boost and cannot
buck the output voltage levels. A solution to its inability to buck, is achieved by adding a
DC-DC convert to the voltage source. This solution, however, comes at the cost of

additional devices, added complexity, reduced reliability, and reduced efficiency.

2.3 Z—source Inverter

L1

A A

e e e

Vc+
: C] — C2 E
l J t} e
W* ' '
L2

 

 

 

 

 

 

Figure 23 Z-source inverter

A typical Z-Source Inverter is shown in Figure 23. Notice that the Z-Source
Inverter uses an impedance network on the DC Link where as the VSI used just a voltage
source (such as a capacitor) and the CSI uses a current source (such as an inductor). This
impedance network is composed of both inductors and capacitors. As such, it offers a

unique combination of the ability to both buck and boost voltage levels to the output

34

unlike the VSI or the CSI. The Z-source inverter can be used for many applications,
including dc-ac, ac-dc, ac-ac, dc-dc power conversions, with any ac or dc source and
load. This work will discuss the Z-Source Inverter as it applies to the SHEV bus.

Therefore, the focus will be on the dc-ac conversion.

Control of the Z81 differs from the traditional PWM for VSIs. Traditional PWM
uses 8 different states (also called vectors). The first six vectors occur when the voltage
of the DC link in impressed across the load. These are also called active vectors or states.
Some examples are:

State 100: top switch of the phase A leg is closed

bottom switch of phase B leg is closed
bottom switch of phase C leg is closed

State 100 implies that the top phase A switch (Sap) is closed and that the top
switches for phase B and C are open. This also implies that the corresponding bottom
switches are of the opposite (or complimentary) state. See Figure 24 for a schematic

example of the 100 state.

35

 

\W

ll

 

 

 

 

 

l

 

 

0

.I

 

 

 

0

éan

sbn

3.69
W
i VTM
.l
éCn

Figure 24 Illustration of the 100 switching vector

Figure 25 Shows the 01 1 vector.

 

 

 

 

 

 

 

 

 

g 3.39 $199 5,09
i K I .l
J
i: : W
lVd
l W
l ‘ VTM
l
0 \ K
- . l l 1.
San Sbn SCn

Figure 25 Illustration of the 011 switching vector

36

As mentioned above, there are a total of six active states:

° V001
' V010
° V011
' V100
' V101
’ V110

The VSI also has two open circuit zero states in which the load terminals are shorted

together through either the top dc rail or the bottom rail (vector V000 or V1”). See Figure

 

 

 

 

 

 

 

 

 

 

26.
r 5.39 $139 $39
- I
l ,l l l
l
x l W
(W W
‘ VTM
k x. k
- ., l l 1
San Sbn SCn

Figure 26 Illustration of the V1" switching vector

This is called the open circuit zero state because there is no complete circuit between the

DC Link and the load (open circuit). Remember that the VSI does not allow the short

37

circuit zero state because this would cause a short circuit across the voltage source
leading to extremely high currents through the switching devices resulting in their

destruction. See figure 27 for a diagram of the traditional PWM wave forms.

 

Figure 27 Traditional PWM Control Method

In this diagram, Va*, Vb*, and Vc* are the reference sinusoidal voltage signals. In this
diagram, the frequency of Va*, Vb*, and Vc* are << the frequency of the triangle carrier
wave and Va“, Vb*, and Vc* appear as straight lines during this short time period. The
switching mechanism (modulator), as discussed in chapter one closes the related upper
switch when its reference voltage signal is > the carrier wave and opens the switch when
the reference voltage signal is below the carrier wave. In the lower portion of this
diagram, the switching signals are shown. Logic state high corresponds to closing the
switch and logic state low corresponds to opening the switch. Also notice the open

circuit zero state (e. g. V000) switching signals.

38

2.3.1 ZSI Simple Control
The ZSI has all of the above 8 switching vectors plus three additional short circuit zero
vectors. These short circuit zero vectors occur when both the upper and lower switch of a
single phase leg are closed (e.g. Sap and San are both closed, or both pr and Sbn, or
both Scp and Sen are closed). In the case of the ZSI, this short circuit condition is
acceptable due to the inductors of the impedance network which limit the current levels.
To summarize, the ZSI allows all three types of states. It allows all six of the active
states, both of the open circuit zero states, and all three of the short circuit zero states.
Each of these state types has a specific purpose in the control of the ZSI and they will be

discussed below.
2.3.1.1 Active States

During the 1,0,0 active state, the line to line voltage across phases AB of the load are
equal to the DC Link voltage, also phase AC is equal to the same. The active states are

used to apply a three phase balanced PWM sinusoidal wave form to the load.
2.3.1.2 Short Circuit Zero State

The short circuit zero state of the Z-Source also known as the shoot-through state [9]

results in the following circuit. During this mode, the inductor current is directly

1 .
charging the capacitors thereby boosting their voltage VC = VCO + Ejlcdt . Since the

39

inductor voltage on average is zero and since it is the series combination of the capacitor
and the inductor that is in parallel with the inverter during the active state, this shoot-
through state boosts the output voltage of the inverter. The amount of voltage boost fi'om

the ZSI is controlled by controlling the amount of time spent in this mode.

Tum—

 

 

 

._I:7’T;:l

Figure 28 ZSI During SCZ State

 

See the wave forms in figure 29 to see the insertion of shoot-through states.

 

Shoot through
zero states

Figure 29 Modified PWM with shoot through zero states [16]

40

 

the sr
isgrc
be ICE
1000 i
conu
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Open
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“her

Shoot-through states must be inserted during periods of time when one or both of
the switches on a phase leg are open. A switch being open occurs when the carrier wave
is greater than the reference voltage wave. Therefore, the shoot-through duty ratio must
be less than the active duty ratio meaning that the modulation index (M) must be less than
100% to use shoot-through. The maximum shoot-through duty ratio of the simple boost
control is limited to (l-M) [17]. The thick curve in figure 30 shows the maximum
obtainable voltage gain, MB (where B is the boost factor) versus M, which shows zero

voltage boost and zero voltage gain at M=l.

: __(T1 {713) Where T1 is the non-shoot-through time period

To is the shoot-through time period

T is the inverter switching period
The shaded area below and to the left of the bold curve is the possible regions of
operation. Notice that as M approaches 1/2, and as B is made to be large, that the possible
gain (MB) grows very large. From the equation above, it can be seen that increasing To
(the shoot-through time period) will increase the boost factor B. It should be noted
though, that the voltage stress across the devices increases as the modulation index goes

down. Therefore, the best combination of M, B, and device ratings must be coordinated

when developing the inverter design.

41

 

DJ -l>- Ur

Voltage gain (MB)
[\J

 

 

 

 

0 0.2 01.4 0.6 V 0.8} 71
Modulation index (M)

Figure 30 Voltage gain of the simple boost control [17]

2.3.1.3 Open Circuit Zero State

During this mode, the voltage across the inductors is much lower (approx. zero) than
during the other modes. Current through the inductor flows into the capacitor increasing

the voltage of the capacitor. As a result, the current through the inductor decreases

IL = 10 + J'VL dt and the output voltage to the load will also be decreased. The extent

to which the output voltage is to be bucked is controlled by controlling the duration of the

42

time period of this mode.

 

 

I [M
C2

_—

 

—
'—

 

Figure 31 Z-Source Inverter in Open Circuit Zero Mode

The Z-source inverter advantageously utilizes the shoot-through states to boost
the dc bus voltage by gating on both the upper and lower switches of a phase leg. Figure
29 shows the schematic of the Z-source network during the shoot-through state.

Obviously the inductor current is boosted during this state. This is because the current

ramps up through the inductors, whose currents are [L = [0 + I VL dt . During the open

circuit zero state, none of the voltages (VC or VL) are impressed across the load.
Therefore the average output voltage goes down as the open circuit Zero state increases.
Therefore, the Z-source inverter can buck (by lowering the modulation index which
increases the open circuit zero state) and boost voltage by increasing the shoot-through
zero state (which increases B). The ZSI can buck, and boost voltage with one simple
efficient structure. The inverter will not be damaged by either open circuit zero states or

by short circuits zero (shoot-through) states and is therefore more reliable. During the

43

active states, the voltages are impressed across the load. Recall figures 24 & 25. With

the ZSI, the switching period is broken down into three separate states:

Open circuit zero state
Shoot-through (short circuit zero state)

Active state

Their respective time periods are T1_Open, To, and Tl_active- And as shown above, the

buck or boost is controlled by controlling these time periods.

2.3.2 Maximum Boost Control

For a given modulation index, this methods achieves the maximum voltage boost
[17]. This method is very similar to the simple PWM control method. In this case
however, all non-active states are shoot-through (boost) states. All six active states are
unchanged. Thus the inverter’s output will not be distorted. There are no open circuit
zero states being used -- only active states and shoot-through zero states. As such, the

boost factor B has changed as compared with the B from the simple PWM method. For

1
To . The PWM waveforms of this method are
1—2—
T

B:

the maximum boost case,

shown below in figure 32.

44

Va
l

 

CD
° / \‘1\/

/ 3! D-

..<

\ / D»
Q

l

7“

 

 

 

8 RI" I
°° UNI—ll
San—_Lfl_fl_ LF

fl
_FILJLIT—LFII Lil]

 

 

 

 

 

:rgC
':

 

 

 

 

 

 

 

 

 

 

 

 

 

 

 

 

 

 

 

 

 

 

 

 

5
'3
I:
I
:1
LL 4 r
j
nC
[—

Figure 32 Maximum boost control strategy [17]

To achieve a given gain, the modulation index does not need to be reduced as
much with this control method as it would with the simple PWM control method.
Therefore, the voltage stress across the switching devices is less with this method. See

figure 33. which shows the modulation index versus voltage gain for this method.

45

 

____________

A

_______________________________________________

b)

    

____________________________________________

Voltage Gain (MB)
N

 

 

p—A
I
l
I
|
I
|
I
I
l
I

__--_---.al_...-—-..---
I
l
l
l
I
I
I
I

at
I
I
I
I
I
I
I
I
I
r
I
I
l
|
I
I
I
I
I
_~__a___-r_---_a_-

4L ll 1

0 0.2 0.4 l 0.
Modulation Index (M)

 

.9
00
p—r

Figure 33 Voltage gain of the maximum boost control strategy [17]

Notice that a gain (MB) of five is achieved with a modulation index of
approximately 0.7 where as a modulation index of .55 was required before.

This method may also utilize third harmonic injection to expand the allowed
range of modulation indexes above 1. The related waveforms are shown in figure 34

below.

46

 

 

 

 

 

 

 

 

.5”
\ “
(Aging
\

 

_I<l

 

 

 

LJ
_IlJ—TL

Fl
Ff Tm

 

Lg'L

 

 

_I
4mm.
1

 

 

 

 

 

 

 

 

 

 

 

 

 

 

 

 

 

 

 

 

 

 

 

 

 

L LgL
d LI—IL

'33

Figure 34 Maximum boost control with third harmonic injection strategy [17]

For the simple PWM method and for the Maximum boost PWM method, the

maximum modulation index allowed is an index of one. In this control, the maximum

modulation index M = — can be achieved at 1/6 third harmonic injection [17]. That is,

.5

the amplitude of the third harmonics injected is 1/6 of the amplitude of the reference
signals (Va, Vb, Vc). The voltage gain is identical to the previous control method for the

same modulation index. That is, the slope of the voltage gain vs. modulation index curve

47

is not changed with the third harmonic injection. The only change is that the curve now

extends further to the right showing that the modulation index now allowed is greater.

 

  
    

See figure 35 below.
: r l
5 ....... 4 ....... : ....... ' ................. a
l I I
t l :I .
E E l ‘
l l I
a 4 ““““ 7 ““““ r """ ; """"""""" i
l l
2 z z :1
V |
a : ; II
'3 3 ------- 4 ——————— : ——————— t —————————————————— 7
on i : I
g, z : .I
_g I I I l
«a 2.______J _______ [___-__-_I _______ I ___________ _
0 ; : '- l
> I I I I l
l I l I
l I l l
l I I I l
1 ....... 1 _______ : _______ .l _______ L ...... I _____
: I l I ' z
I I I I' I"
I : 2| : :
0 t : I, : :4

 

 

 

h—IT

0 0.2 0.4 01.6 0.8
Modulation index

Figure 35 Voltage gain of the maximum boost control with third harmonic injection

[17]

2.3.3 Maximum Constant Boost Control
From the Maximum Boost control method and as can be seen from figure 32, the shoot-

through period varies throughout the reference voltage’s cycle. This causes capacitor
voltage ripple and inductor current ripple that is associated with the output frequency.

The ripple is most predominant at low output frequencies. To mitigate this, the inductors

48

and capacitors of the ZSI’s impedance network must be oversized. A method has been
developed to achieve the same gain but with constant boost through out the output
voltage cycle. This method known as “Maximum Constant Boost Control” [18] causes
less voltage and current ripple and therefore requires smaller inductors and capacitors.

The control scheme is shown in figure 36.

 

mI34

  
   

ff 1;”... r“ ,_-- i
I
a

_JLJ
_I l_
_J_J

 

   

 

 

 

 

 

 

 

 

 

]_l
_J
__J _Jl_l_I]_l"l_F_ l
I] _J J

Ir— ‘LI _I“ ]__l'l_l""
S“ _r _lJ—LI IJ'LI'TI

Figure 36 Maximum constant boost control strategy [18]

_IJJ

 

 

|;_I__l_l

 

 

_l_l_J
I:
:l

 

l

 

 

 

 

 

 

 

 

 

 

 

 

 

 

 

 

 

 

 

 

 

 

 

Notice that there are five reference voltage curves in the diagram above. There are the

usual three sinusoids Vail, Vb‘, Vct separated by 1200 which represent a three phase

balanced system. In addition, there are two shoot-through envelope signals, Vp and Vn

with a periodicity of three times the fundamental frequency (One in the positive portion
of the graph and one in the negative portion). When the carrier signal is above the upper

shoot-through envelope signal, shoot-through of the inverter switches will be triggered

49

and when the carrier signal is below the lower shoot-through envelope signal, a shoot-
through will also occur. By looking at the graph of the five signals, it can be seen that the

shoot-through periods are consistent unlike that of the maximum boost control method.

In the graph above, the distance between Vp and Vn is always J3M and the shoot-

T0 2—J§M ~/§M
through duty ratio averages to — = — = 1" — Where
T 2 2
To is the shoot-through time period
T is the switching period
M is the modulation index

The duty ratio averaged over one switching period of the carrier signal is equal to that of

all other switching periods of the carrier signal.

The goal of this method is to achieve maximum boost while remaining constant. See the

graph of voltage gain (MB) vs. modulation index in figure 37. Gain approaches infinity

when M decreases to —3—3.

50

 

~I [5“!
II

Obsen

and tha
through

this Ihlrc

 

 

 

Voltage gain (MB)

 

 

 

 

 

0.5 0.6 0.7 0.8 0.9 1
Modulation index (M)

Figure 37 Voltage gain of the maximum constant boost control strategy [18]

 

1
As seen before with other PWM control methods, B : T . and since
1—2—0
T
—: _ ,Bcanbeex ressedas =—_.
T 2 p J3M —1

Observe that Vp and Vn are both third harmonics of the reference voltages Va'i, Vb*, VJ,

and that Vp and Vn ride along the top and bottom of the reference voltages. Shoot-
through switching is controlled by comparing Vp and Vn to the carrier signal. Suppose

this third harmonic wave form were added directly to the reference voltages.

51

 

 

rt “Ii
- , >520

’ V V I

Figure 38 Third Harmonic Injection into Reference Signals

 

And then instead of the Vp and Vn envelope signals being sinusoids, we replace them
with straight lines. What has been done is that the third harmonic component has been
subtracted from the envelope signals and has been added to the reference signals (third
harmonic injection). This leaves the reference signals shown above and now the

envelope signals are straight lines as shown below.

52

 

1‘

Sap.

pr1
Sep'

Saul

Sbnu
Scnl

Fig
The rest

“Wrmj

 

 

 

 

 

 

 

 

V‘ i . 5Vn

 

 

 

 

SapUJ .=

prLJ—__,__T_ ‘ .1
3°“ | g LF_J_—il_l_LT-¥i
Saran—r .LL‘_’1_'_["_J*
Sbnll i g T L If L!__F’
San I I I. ._Lfl_l ? i’

Figure 39 Third Harmonic Injection with Max. Constant Boost PWM [18]
The result is the same maximum constant boost PWM as before and the shoot-through

 

 

_J—

 

 

 

 

 

 

 

 

 

 

 

 

 

 

 

 

 

 

. To fiM
duty ratio is the same as before, ?‘ = 1“ T , and the gain equation is the same as
13 M
before V 0 _ MB — — G Where
dc 2 «EM —1

‘30 is the output peak phase voltage

53

V [C is the input dc voltage

(

G is the gain (MB)
The difference is that with this control method, the range of M is increased to §J§ . The

voltage gain vs. M is shown in Figure 40. The voltage gain can be varied from infinity to

. . 3 2 . . .
zero smoothly increasmg M from -—3— to — wrth shoot-through states (solid curve 1n

J3

Fig. 40) and then decreasing M to zero without shoot-through states (dotted curve in Fig.

40).

Voltage gain (MB)

d’

 

I —
-’,'
—d

O
0 0.2 0.4 0.6 0.8 1 1.2
Modulation index (M)

Figure 40 Voltage gain of the maximum constant boost control with third harmonic

injection [18]

V =BVdc =(I/EG—1)Vdc, A

Voltage stress across the switching devices is s

graphical comparison of the voltage stresses of the control methods is shown in figure 41.

54

 

 

9 1 l I
——|— simple control :
. g..— mam'mmn boost

—6— max. const. boost

 

 

\l

 

Voltage stress] Vdc

 

 

 

 

Voltage gain

Figure 41 Voltage stress comparison of different control methods [18]

As can be seen, maximum constant boost causes higher voltage stress than maximum

boost; however, maximum constant boost eliminates the output frequency ripple.

2.4 Input/Output Equations of the Z-source Inverter

The function of the Z-Source Inverter is to convert a DC input voltage into an ac

output voltage. Using V0 as the DC input voltage and vi as the DC Link voltage across

55

the inverter, 13,- : B * V0 [16]. From this it is seen that the DC Link voltage is

1
controlled by the boost factor B, where B = ———T—- 2 l .
0

1—2——
T

Where
To is the shoot-through period
T is the switching period (period of

the carrier signal)

The inverter switches deliver this DC voltage to the output using PWM to convert the DC

to a sinusoidal ac voltage. The output of the inverter is
i}. V
,. _ a: _l_ _. >1: a: 0
vac —M —M B
2 2
Where

vac is the peak ac output phase voltage

M is the modulation index.

This equation shows that the output can be bucked or boosted by choosing the

correct buck/boost factor BB where B B = M * B = (0 ~ 00) .

Where M is the modulation index. This equation shows that the output voltage can be

Stepped up and down by choosing an appropriate buck—boost factor, B B.

BB=M*B=(0~oo).

56

.f' L".-.._n.n.¢'

If.

3. INTRODUCTION TO POWER ELECTRONICS
TOPOLOGIES AND THEIR OPERATING PRINCIPLES

The purpose of this thesis is to investigate traction drive systems for a series hybrid
electric bus. The purpose of the power electronics is to control the flow of power from or
to the ICE’s generator, to or from the batteries, and to or from the traction motor. The
voltage must be managed in terms of ac frequency and magnitude. The amount of power
flow to or from the system batteries must be managed to control the state of charge of the
batteries. Power flow to or from the traction motor must be controlled to provide the
desired acceleration/deceleration and speed of the bus. In this process, of controlling the
power flow for the battery, traction motor, and generator, all must be managed so that
optimal efficiency is achieved. Several circuit topologies have been chosen for further
investigation. Below is a summary of the topologies which will be discussed in more

detail.

0 Traditional Input Rectifier/V SI to Output VSI

0 Input Rectifier/V SI to Output CSI with Z-Source impedance network
(battery in parallel with Z’s capacitor)

0 Input Rectifier/CSI to Output VSI with Z-Source impedance network
(battery in parallel with Z’s capacitor)

0 Input RectifierN SI to Output CSI with Z-Source impedance network

(battery in parallel with Input Converter)

57

Each topology and its operating principles will be discussed. The traditional Input
Rectifier/V SI to Output VSI will be discussed first due to its historical familiarity. This
topology will be used to discuss the general function of the power electronics to deliver
and control power. Chapter two discussed the specific methods used to control the
inverters. This chapter discusses not how it is controlled (SPWM, etc.), but what it is
being controlled to do. Where power is to come from and where it is to be delivered to.
The hybrid bus has the flexibility of choosing where power is to come from and where it
is to be delivered to. The goal is to make choices that optimize efficiency. In general,
this means that the ICE and generator are to be operated at only one or two power and
angular velocity settings to supply average power required. The batteries are to provide
the “dynamic” load. That is the power above and below the average power needed. In a
typical bus route, the bus must accelerate and decelerate many times. The total energy
expended by the bus on its bus route divided by the amount of time it takes, gives us the
average power. In this example, while the bus is accelerating, the traction motors
typically requires more power that this average power level. This additional momentary
power variance is to be provided by the batteries. With this method, the ICE power vs.
angular velocity is chosen to be its most efficient operating point. Operation at the most
efficient point is not possible with the conventional bus since the ICE alone must provide
all of the range of angular velocity and of power that the bus’s wheels require. Also, the
hybrid bus is capable of recapturing deceleration energy which is wasted by conventional
busses. To move power consistent with these goals, several operation modes are

encountered. These modes are listed here:

58

Hybrid Operational Modes:

1. Battery and Traction Motor mode
The traction motor is lightly loaded, the battery is supplying power
to the traction motor, and the generator is off.

2. Battery, Starter Motor and Traction Motor mode
The battery is supplying power to the traction motor and
simultaneously providing power to the generator. The generator is
momentarily being used as a motor to start the internal combustion
engine (ICE).

3. Generator, Battery Discharge and Traction Motor mode
The traction motor is medium to heavily loaded. The generator
and the battery are both supplying power.

4. Generator, Battery Charging and Traction Motor mode
During this mode, the Traction Motor load has decreased and the
battery transitions from discharging to charging.

5. Regenerative Braking and Battery Charging mode
During this mode, the load at the traction motor becomes
negative and the traction motor functions as a regenerative

brake (generator). The battery is charging.

In order to be considered for use on the hybrid bus, a specific power electronics circuit

topology must be able to function in each mode. There are four topologies to be

discussed. The first is the traditional Input Rectifier/V SI to Output VSI shown below.

59

This circuit architecture uses a capacitor in the center to provide constant voltage and
varying current. Because of the constant voltage provided to the inverters, and the
bidirectional current flow in the inverters, these inverters are called voltage source

inverters (VSI).

60

P
generative
M Brake
LT RT
480 V
-100 to
+250 kw

Traction
Motor I Re-

 

 

 

 

 

 

 

 

 

 

l; '.=
34:2 ~ Tee
_1 —' (D
h N
e. .. .g
g @4111 QiJ—fi 1:
g (D fi fl 0) 3
'fi fl
8 E .5
~ “4:2 as»-
'5
< > §
II
I—fiV
U
I.
jII.

Cs

Source Inverter
sas-p SPs-p Ses-p
l3}: lg}: Jig}

be
1K}L K? it?
sas-n ébs-n SOs-n

 

 

 

 

’as

 

st = 5 kHz

 

 

 

I Starter mfm—I
Ls
Mme

Motor Rs

0
>

Generator
360 to 480 V
0 & 120 Hz
0 to 150 kw

Figure 42 Traditional Input Rectifier/VSI to Output VSI. The left
end of the schematic shows the generator which is connected to the
ICE. This end is considered the power flow input. The term rectifier
is also used since the input converter may be used as a passive

rectifier. Voltage sources in the center and a VSI at the output

61

y g? -=.-¥-.-r+*.q

The second circuit architecture is the Input Rectifier/V SI to Output C SI with Z-Source
impedance network (battery in parallel with Z’s capacitor). Because of the bidirectional
current flow allowed by the input converter (on the left of the schematic), the input
converter is called a VSI. Notice that the output converter (right side) only allows one
directional current flow but bidirectional voltage blocking. As such, it is called a CSI.
The network in the center is made up of the unique combination of capacitors and

inductors and is called the Z-Source (Impedance-Source).

62

 

 

 

 

 

 

 

 

 

 

 

 

 

 

 

 

 

 

 

 

 

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8:8? > 3
233m >N
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3 3 7e; 5‘“
I'— ‘— r it?
a: (D
it"
I ljSI i...
gt: +5- $3
ll ,
m
0
[Fe—Q gig—.3
h 0
3 e33 :9 “é
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C31 ‘ b“ 3
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3
& >£§
gag §§§
Egg g §g§

Figure 43 Input Rectifier/VSI to Output CSI with Z-Source impedance network
(battery in parallel with Z’s capacitor). The left end of the schematic shows the
generator which is connected to the ICE. This end is considered the power flow
input. The term rectifier is also used since the input converter may be used as a

passive rectifier. Z-Source network in the center and a CSI at the output

63

The third circuit architecture is the Input Rectifier/C SI to Output VSI with Z-Source
impedance network (battery in parallel with Z’s capacitor). This topology is the same as
directly above accept that the converters are swapped end for end. In this case, the input
converter is the C SI which allows only one directional current flow and bi-directional
voltage blocking while the output converter is the VSI converter. Its schematic is now

shown below.

64

 

o E
$5» >- 5
‘8'“ 8
3.32 >N
3cm §+
a?» ,_ 2

.— I
_r

SIT-n

 

 

 

A

Traction Inverter
S‘IT p SZT-p S3T p
g} 4)}? @ Traction

 

 

r

sm sin.
st= 10 kHz

 

 

 

 

L2+

 

 

 

 

 

 

 

 

 

 

 

 

 

._ 93-154 r";

g m a)

g 8 s:

E ?-. ‘ .
g as, 3" 1“ 5"; e
‘3 £1 ‘6 ‘ g E
0’ $4 «He—rem ¥ ”a.”

3 e

> z

2.3? SE 33%:

egg :3;

Eco: ‘>-9 n°°

Figure 44 Input Rectifier/CSI to Output VSI with Z-Source impedance network
(battery in parallel with Z’s capacitor). The left end of the schematic shows the
generator which is connected to the ICE. This end is considered the power flow
input. The term rectifier is also used since the input converter may be used as a

passive rectifier. Z-Source network in the center and a VSI at the output

65

The fourth circuit architecture is the Input Rectifier/VSI to Output CSI with Z—Source
impedance network (battery in parallel with Input Converter). This is like the second

circuit shown above except that the battery position is different. The schematic is shown

below.

66

generative
Vm

me Brake
Rr

I

Traction

Motor/ Re-
480 V
-100 to +250 kW

 

Lr

 

L
I
I
1
I

 

 

 

 

 

Traction Inverter

'

 

SIT-P SZT-P 3.3T-P
5.
PI 13L PI
S1T-n SIT-n S3T-n
st= 10 kHz

 

v

 

 

”66‘

+ L1

+ +
‘c1 C2"
”615‘

- L2 +

 

I

Cs

r e I; r
5: st .1?

V

 

 

Source Inverter

 

 

V

 

 

qu-p Sbs;p SCs-p

 

FSW = 5 kHz

360 to 480 V
0 8: 120 Hz
0 to 150 kW

Generator
I Starter
Motor

(5
>

Figure 45 Input Rectifier/V SI to Output CSI with Z-Source impedance network
(battery in parallel with Input Converter). The left end of the schematic shows the
generator which is connected to the ICE. This end is considered the power flow
input. The term rectifier is also used since the input converter may be used as a

passive rectifier. Z-Source network in the center and a CSI at the output

67

Sections 3.1 to 3.4 discuss the operational principles of these four circuits. Because of its
historical familiarity, the traditional VSI — VSI topology example (section 3.1) will be
used to explain the operational modes in detail. Also, a discussion regarding the pros and
cons of using a high voltage battery vs. a low voltage battery is included in this section

(section 3.1.3).

3.1. Traditional Input Rectifier/V SI to Output VSI
The schematic of this topology is shown above in Fig. 42 Recall from chapter 2, that the

VSI must have a voltage source that is greater than the desired output voltage level.

J5 Vdc

VLL :_ M

J52

Where: VLL is the line to line RMS output voltage

 

Vdc is the dc input voltage

M is the modulation index [ 0 S M S 1.0 ]
Therefore, the VSI is considered a buck converter. If Vdc cannot be as high as we would
like, a DC/DC converter can be added to the circuit architecture to achieve the required
input voltage. First, this circuit’s functionality in each of the five modes will be
discussed. Following that, the use of a high voltage battery will be discussed and then the

use of a low voltage battery with DC/DC converter will be discussed.

3.1.1 Battery & Traction Motor Mode

In the first operation mode, (Battery and Traction. Motor) the entire load is being

68

supplied with battery power though the traction motor. This mode assumes that the state
of charge (SOC) of the batteries has reached the maximum allowed for longevity.
Therefore, the ICE and generator will be off and all switches of the input
VSI will be open. Current from the battery will flow to the traction motor through the
output VSI. The output VSI will control the switching functions to provide the required
voltage level and frequency. This mode is typically encountered when the bus is sitting
still and the bus driver has depressed the accelerator pedal so that the bus will begin to
move. In this particular state, a low voltage and low frequency output is required from
the output VSI. The low voltage is controlled by using a small modulation index “M”
and the low frequency is controlled by a motor controller that is designed for permanent
magnet alternating current motors (PMAC). The motor control algorithm controls the
voltage phase to provide “(1” and “q” axis currents such that the stator field is 900 out of
phase with the rotor field, i.e. the “d” axis current = 0. This efficiently accelerates the

bus with the maximum torque that is available at the chosen current level.

3.1.2 Battery, Starter Motor, Traction Motor Mode
The second mode is the “Battery, Starter Motor and Traction Motor mode.” Typically,
this mode follows the first mode discussed. For example, when the amount of power to
the traction motor that is requested by the bus driver exceeds the average power level, or
the SOC of the battery is below the desired level, the ICE should be started. In this state,
the battery is providing power to the traction motor and to the generator. The generator is
now being used as a starting motor for the ICE. The switching of the input VSI is now

being operated to provide the desired starter motor speed (output voltage frequency) and

69

voltage level to produce the needed starter motor torque. As mentioned before, the
voltage phase is controlled to provide the best torque angle. Once the ICE is started, the
input VSI will begin active rectifier mode. It is at this moment that the generator stops
functioning as a motor and begins functioning as a generator. The modulation index and
VSI phase angle vs. generator phase angle are controlled to pull the average power from
the generator. It is the input VSI that controls the power of the ICE. The ICE is
governed to maintain the specified angular velocity. The following example helps to

explain this power flow control.

Power flow from the generator to the DC link can be modeled as follows.

Generator
Inductance

AA/‘K/‘R
.r’ ‘v‘ ‘r V "L

+ i / + Input
Generator VS'
/\/ EMF /\/

 

 

Figure 46 A single phase model of the Voltage Source Inverter and the Generator

In this model, the sinusoidal voltage source on the right represents the voltage of the VSI
and the sinusoidal source on the left represents the EMF voltage of the generator.

Between these two sources is the generator’s inductance and negligible resistance. Power

70

flow through this inductance can be seen to be

_ VVS] Simgm) _ VGEN Sin(6(}EN)
¢ _
Z

Shifting the phase of both sources by 900 does not change the power. We then have:

 

P1

: VVSI COS(6VS] ) _ VGEN COS(BGEN )
¢ Z If we let VGEN be the

 

P1

reference voltage and treat the angle as 0 we arrive at the following.

_ VVSI COS(6VSI) _ VGEN

(15 — Z In this equation it can be seen that the variables

P1

 

that are controlled by the VSI are the voltage magnitude and angle of the inverter. VGEN
is determined by the angular velocity of the generator which is determined by the angular
velocity of the ICE, Z is a constant of the generator. Accordingly, power flow (positive
or negative) can be controlled by controlling the Vvs. and 9V3] relative to the voltage

magnitude and angle of the generator.

3.1.3 Generator, Battery Discharge, Traction Motor Mode
The third mode is the “Generator, Battery Discharge and Traction Motor mode.” In this
mode, the input VSI operates as an active rectifier. The voltage and phase angle of the
generator is monitored. The modulation index and VSI phase angle vs. the generator
phase angle are controlled by the VSI to pull the specified power from the generator.
Generator voltage polarity matches that of the DC link, current flows through the
rectifier, and power is thereby transferred to the DC link. In this mode, the power

requested by the bus driver exceeds the average power which the ICE and generator are

71

now providing, and the excess power is provided by the battery.

3.1.4 Generator, Battery Charging, Traction Motor Mode
The next mode is the “Generator, Battery Charging and Traction Motor.” Very similar to
the mode discussed just above. In this mode, however, the battery is now receiving
charge rather than providing power. This mode is typically entered when traction motor
power drops below the average power level. When the bus driver raises the accelerator
pedal, the output VSI reduces modulation index (thereby voltage) and or phase angle to
decrease the amount of power being pushed to the traction motor. If this amount of
power is below the average power being supplied by the ICE, generator and input VSI,

the excess power flows into the battery.

3.1.5 Regenerative Braking and Battery Charging Mode
The “Regenerative Braking and Battery Charging mode” occurs when the traction motor
becomes a generator. If the bus is moving and the bus driver allows the accelerator pedal
to return to the full up position, the output VSI will enter the regenerative brake mode.
This is accomplished by control of the Traction Motor VSI phase angle and modulation
index. By causing the output VSI phase angle to lag that of the traction motor, the
traction motor pushes power to the DC link through the output inverter thus functioning
as a generator (regenerative brake). See Figure 47 below. The negative torque slows
the bus and generates power that will be stored in the battery. In this mode, the ICE may

be stopped or may continuing running depending upon the needs of the battery SOC and

72

the allowed max rate of charge. When power flow is controlled via inverter phase angle
vs. phase angle of an ac machine, voltage magnitude is also controlled via modulation

index. With these two controls, power factor can be maintained near unity.

Control of the transition from motoring mode to braking mode can be shown with the
following graphs. While in motoring mode, the phase angle of the output VSI is

controlled to lead that of the traction motor. See Figure 47 below.

 

 

 

 

 

 

 

 

 

400-” . I T I Battery current
zooo‘ ---------- e ---------- 1- -—-+——:— % --~--
(A) 0 I ---------- I ---------- I ---------- I --------- 2- ---------
-200.0~ --------- Z ---------- Z ---------- I ---------
400.0
1 0 . . . . Traction motor
' - - ' - ”Mamet!!! ..........
/. x . {.75. . /'-\
4' ‘ . 5' \. . /.v .
/.-/'"' it. .44 \- . //,.-" . ”I, Traction Inverter
(v) 0.0«l - k.4" ------ “It. ------- [4." ------- \I -I ----- / 4:5 - - 2- - - - X»:- - - - Vret'
/-' ' (a. /-' - {3: z - Ia. ——————————
f! \\\‘7// \\‘_../ \\\..,
1'0 r I I i I 1
NS)

Figure 47 Voltage Phase Angle Control

As can be seen, the battery current is positive showing that the battery is discharging
(providing power to the traction motor). The transaction to the regenerative mode is
effected by retarding the phase angle of the VSI with respect to the angle of the traction

motor (brake).

73

400.0

 

 

 

 

 

 

 

 

 

 

 

Battery current
200.04 -----------------------------------------------------
(A) o .. .............. ..................................
-2oo.o —'—. - -222- a
400.0 -------------------------------------------------
-600.0
1.0 . . . Traction motor
: ' : ' "519-09.931.31 ..........
_/ . ______ \ ___________ (f raconnve er
(V) 0.0“ ' 1‘5,” """" 3 1 ~33 ; 3 M" Vrel'
/;/ . l \I“\ i // ——————————
1-0 I I I T i I I I I I
III»)

Figure 48 Voltage Phase Angle Control

The above graphs shows the case where the ICE & generator are off, and the traction

motor voltage leads the inverter voltage thus pushing power from the traction motor (now

in regenerative mode) to the DC link through the output VSI. the negative battery current

in the top graph shows that the battery is being charged by the regenerative process. The

VSI’s reference control voltage is shown in the graph above since it more clearly

illustrates the principle than the actual discrete voltage pulses of the PWM waveform.

Voltage magnitudes are normalized.

74

3.1.6 High voltage battery

Figure 42 above shows the high voltage battery option where the battery is in parallel

with the capacitor. In this case, the battery will need to be at least

275 480
V, =

C J3

In order to achieve a sinusoidal voltage of 480 VRMS line to line. This calls for a series

2 784Vdc

 

combination of at least 62 batteries if we use the 12.8V lithium ion battery as an example.

3.1.7 Low voltage battery with DC/DC converter
For the low voltage (LV) battery case, the battery will require a DC/DC converter
between the LV battery and the DC link. If a 205 V battery is used, this would be 16
12.8 V batteries in series. The DC/DC converter must be able to boost voltage when
delivering power from the batteries to the DC link and it must be able to buck voltage
when delivering power from the DC link to the batteries. In this case also, a minimum
voltage of 784V at the DC Link must be achieved which requires a boost gain of 3.8.

Below is a schematic of a typical DC/DC converter.

75

fl *—

 

 

 

 

 

 

 

 

 

 

Boost DC-DC With Buck
Battery
'33“ —' Storage
(+) V. (-) /
--’b"6‘ O
81 (+)
(+) ___ sz \ 7: Va DC-Link
VBATT (-)
(-l
0
Figure 49 DC/DC Converter

The typical DC/DC converter for such an application has the following gain equation:

The switch function is denoted by the switching period T where T=T0N + Topp. For the
inductor, the average voltage across the inductor during steady state is zero. This gives:

: (VIN *TON +(VDC —VIN)T0FF) : O
T

SI

VIN *TON : (VDC _ VIN )T OFF
VIN*T0N+VIN*T0FF :VDC*TOFF
VIN(TON+T0FF)=VDC*TOFF

 

 

 

VDC _ T

[N T OFF
VDC _ T
VIN T ‘- TON

Section 3.1.8 below gives a comparison of the high voltage battery versus low voltage

76

and discusses the pros and cons. The comparison is discussed here in the back to back

VSI section, but the principles apply to the other topologies as well.

3.1.8 Comparison of the high voltage battery vs. the low voltage battery

High Voltage Battery Option

The high voltage battery has the advantage of being able to connect directly to the
DC-Link without the use of a DC-DC converter. This option is simple in design
and uses fewer electronic components. It does, however, require a greater number
of batteries to be stacked in series. There is also a fundamental problem with
connecting batteries in series. If the Lithium Ion battery is used as an example,
due to its lighter weight and relatively fast charging rate, e. g. Valence
Technologies U27-FN130, it can receive a maximum charge current of 100A. A
series connection of 51 batteries is needed to achieve a total voltage of 654 volts.
With such a series connection, a failure of any one of the 51 batteries will cause a
serious degradation of performance. The reliability equation for series-parallel

arrays of electronic components is as follows.

R(t) = [1- p"F"(t)]'" -[1 - (1 -qF"(t))l'"

Where:
0 R(t) is the reliability,
0 p is the conditional probability of a short,
0 q is the conditional probability of an open,

77

o m is the number of parallel paths,
0 n is the number of components in series,
0 F(t) is the cumulative distribution function of time to

failure of series components [19]

In this particular case, the short circuit failure rate is quite low for batteries.
Typically, battery failures occur from an increase in resistance and or a decrease
in voltage. As a result, reliability decrease is much more likely to occur from

series connections than from parallel connections.

If it is necessary to receive a charge at greater than 100 A, additional parallel
combinations of 51 series batteries will be needed. As can be seen, the number of
batteries required can add up quickly. Also, with series connections, each battery is
slightly different than the other batteries that it is connected to (different internal
resistance, different voltage level, etc.) As a result, some of the series batteries are
putting out more power than others. Under high current levels, a poorly firnctioning
battery may actually be dissipating more power than it is providing. The net result is a

compromised performance of the whole stack.

78

l0 VOI=V1' onRr

 

V02=V2 ' onRz

 

 

+ ' If (V1m1)< Io< (Vz/Rz).
v0 g RL then

— V01<0, M1 dissipates more
power than it delivers.

 

 

V02>O, M2 provides power

 

Figure 50 Series Combination of Battery Cells; See Notei

As the age of the batteries increase, these cdnditions worsen. Due to these
factors, the high voltage battery has a lower reliability than the DC-DC boost and
low voltage battery combination. Battery management electronics can be added
to monitor individual battery condition and allow for by—passing weak cells, but

this adds to the complication and cost.

Typically, a high voltage series stack is more costly than a low voltage stack of
the same Amp hour rating as it is easier for the manufacturer to assemble large
battery cells than two smaller cells in series. Also with the high voltage battery
stack comes the additional safety concerns for service personnel, rescue workers,

CIC.

 

i Used with permission from Chen, Lihua 04-04-07

79

Low Voltage Battery with DC-DC buck/boost converter

With a low voltage stack, a DC-DC buck/boost converter is required. Since the
low voltage stack voltage level is less than that of the DC Link, a boost converter
is required between the DC Link and the batteries. This boost function is required
when the batteries are delivering power to the DC Link. When the DC Link is
delivering power to the batteries, a buck function is required. The disadvantage
of this low voltage arrangement is that this additional circuit (DC-DC buck/boost
converter) is needed. Along with the additional circuitry, come additional cost
and complexity and lowering of reliability due to the additional components. The
advantages of this design however, is that the total number of batteries in series
will decrease, and this will increase system reliability for the reasons discussed

above.

The use of a DC-DC converter adds an extra level of control for the battery SOC
by controlling the DC output voltage level. Although this level of control is not
needed to control battery SOC since coordinated control of the generator
inverter/rectifier and traction motor inverter/converter also achieve SOC control.
Still, the DC-DC converter can provide this as well as an added level of battery
health monitoring and control. The low voltage option also reduces voltage levels
present when the vehicle is not running which can reduce risk factors to
passengers and emergency personnel in the case of a collision involving the bus.

The low voltage option reduces the energy losses during low power operation by

80

reducing the DC rail voltage and thereby the voltage across the IGBT modules.
THD is reduced during lower power operation by reducing the DC rail voltage to
lower output voltage and thereby keeping the modulation index of the inverters

higher which reduces THD.

Summary of High Voltage Battery Option vs. Low Voltage Battery Option

High Voltage Option
Pros:
0 Battery Voltage = DC Link Voltage and a DC-DC converter is not
Required
Cons:
- Greater number of Batteries in series and a related decrease in reliability
of the battery stack
Low Voltage Option
Pros:
0 Lower number of batteries in series and a related increase in reliability of
the battery stack.

0 Extra level of control for battery SOC

0 Reduced non-operating voltage levels

0 Reduced energy loss during low power operation
0 Reduced THD during low power operation

Cons:
o A DC-DC converter is required
0 Decreased reliability due to additional components.

81

3.2 Input Rectifier/V SI to Output CSI with Z-Source impedance network

(battery in parallel with Z’s capacitor)

The schematic for this architecture is re-displayed here for convenience.

Source Inverter Traction Inverter

Generator! ‘ ‘ ‘ ‘ ”NIX ‘ ‘ ‘ ‘ Traction
Starter Jlé Jig} lg} EI- LIL 5L Motor I Re-
Motor . . ~

 

 

generative
_ _ 99 _ Brake

W»
G

if if I”? a at at

f -fm- - - -

Figure 51 Back to Back VSI, CSI with Z-Source Network

Re-displayed for convenient reference

 

 

 

 

 

 

 

 

 

 

 

 

 

The meaning of the different hybrid operational modes has been well covered in the
previous section. However, some discussion is needed regarding regenerative braking.
In the case of the back to back VSI discussed in section 3.1, the regenerative power flow

is easy to visualize since its current flow can simply reverse directions on the De link.

82

 

Current Flow During Motoring Mode
Current Flow During Regen. Mode

a a a * a a a

VI

VG ,m ‘Wm
TIFF . a «a e

v v v

 

 

 

 

 

 

 

 

 

 

 

 

 

 

 

Figure 52 Back to Back VSI topology

Showing the direction of current flow for regenerative braking and
for motoring modes
In the case of the VSI-Z-C SI shown in Fig. 51 above, the Traction Motor/Regenerative
Brake is driven by a CSI. The CSI uses reverse blocking IGBT modules which only
allow current flow in one direction. Therefore, it is somewhat less obvious how the
regenerative power flow occurs. To go into regenerative braking, the C SI shifts its phase
so that the polarity of the traction motor inverts. The voltage across the stator windings is
phase shified such that the stator field is now resisting the rotation of the rotor. The
traction motor is now a generator with its positive terminal on the bottom rail of the DC

link. See Figure 53.

83

 

Active State (Regenerative Brake Mode)

 

 

 

IL
—>

——+rrnr\
+L1-

 

 

 

+_rji+ +"—_1_+

 

 

 

 

Figure 53 Back to Back VSI, CSI Z-Source Topology

Modeling the active state of the regenerative mode

During the active state of regeneration, the output inverter functions as an active rectifier
and can be modeled as a DC source. During this active state, the regenerative energy is

being stored in the Z—Source inductors. The voltage across each inductor is

V1, = V,- + VC where Vc is the capacitor voltage. This voltage level is greater than

that of the previous state and therefore the inductor magnetic field is building. The next
state is the open circuit zero state for the CSI and active state or short circuit zero state of

the VSI. The short circuit zero state of the VSI is shown below.

84

 

CSI Open Circuit Zero State
, (Regenerative Brake Mode)

IL

——>

 

 

 

 

 

 

W—o
(LEW
VG W

 

 

 

 

 

 

 

Figure 54 VSI-Z-CSI Regenerative Mode with VSI in SCZ

It is during this state that regenerative braking energy is transferred to the Z-Source

capacitors and batteries thereby raising their voltage and SOC.

3.2.1 High voltage battery

The high voltage battery option is shown in the schematic above. The Z-Source topology
has the ability to both buck and boost voltage levels. Therefore, the high voltage battery
does not need to be as high as in the back to back VSI topology. This mitigates some of

the concerns associated with the high voltage case.

85

3.2.2 Low voltage battery with DC/DC converter

The low voltage option is still applicable. With a DC/DC converter connected in parallel
across the Z-Source capacitor, all of the hybrid modes still function in the same manner
as with the high voltage battery. With a DC/DC converter controlling the voltage on the
DC link, an extra degree of control has been added to the overall design. It is necessary
that this now be coordinated with the modulation index of the inverters which adds some

complexity to the controls.

3.3 Input Rectifier/CSI to Output VSI with Z-Source impedance network

(battery in parallel with Z’s capacitor)

The schematic is re-displayed here for convenience.

 

 

 

 

 

Source Inverter Traction Inverter
- # - W L A

16 1K1 1K M} J6}: IS} Traction

Generator I Motor I Re-
Starter Motor _ generative

__ _ ; Brake

J:
VG for? . Vm

 

 

 

 

 

 

 

 

 

    

“i I I

Figure 55 CSI-Z-VSI Topology

-frr‘v

86

In this case, the regenerative braking is again apparent since current direction on the DC
link is allowed to change directions as was the case with the back to back VSI. The
hybrid mode that should be explained here is the ICE starting mode. While the generator
is operating in generation mode, power is transferred from the generator to the DC link.
That is, positive current is delivered to the positive voltage DC link. However, to use the
generator as a starter motor, power needs to be delivered in the opposite direction. The
power direction cannot be changed by simply changing the current direction. The CSI

uses reverse blocking IGBT modules.

<VCSI> : VB and
<VCSI> : ZVBDST + <VCSI_AO >(1 _ DST)
Where:
<Vc51 _ Ao> is average C SI inverter voltage during the

VSI’s active and open circuit states
Ta is VSI active time period

To is VSI open circuit time period
TST is VSI shoot-through period

T=Ta+T0+TST
DST IS TST/T
V — 2V D
V 2 B B ST
< CSI_A0> (l—DST)

 

For the starting operation, the Z-source network will need to be operated with a VSI
short circuit (shoot through) duty cycle large enough to cause voltage inversion. When
the shoot-through state is over on the VSI end and the VSI goes to the open-circuit mode,

the voltage on the inductors L1 and L2 will invert. The magnitude of this inversion must

87

be >VB so that Vcsr becomes negative. During this VSI open circuit state, the CSI

switches will be operated to deliver power to the generator so that it is used as a motor.

Simulation of Reverse Power Flow to Generator

MATLAB CODE
VB=1; %Battery Voltage normalized
T=l; %switching period normalized
Dst_1=0;
Dst_end=.9;
Dst=Dst_1 :(Dst_1+Dst_end)/l00:Dst_end; %shoot-through duty cycle

for i=1:length(Dst)

Vlact(i)= (VB*(1-2*Dst(i)))/(l/T-Dst(i));

end

plot(V 1 act,Dst)

xlabel('DC Voltage across Inverter')
ylabel('Shoot-Through Duty Cycle')

title('Plot of DC Voltage vs. Shoot-Through Duty Cycle')
grid on

88

 

 

 

 

Plot ofDC Voltage vs. Shoot-Through Duty Cycle
0-9 T . I II T. I I I l
as ------- ------ 2 ------ é ------- i ------- ----- —
07 ....... g .............. ...... .5 ............ L ..... _
2 E E E E 5
3.0.5 ------- ------------------- -
Q : : : :
3:? : : : :
8 0-5 ------- ------- a ------- . ------------ -
J: I l I u
U) I I I I I
3 . . . .
.9 0,4 ................... _
J: I I l l I I
I-. = a s a a a =
e 0.3 ------ -—
0° ' : : ' : : : :
0.2 ....... ....... g ....... g ——————— —
0.1 ....... g ............................ ...... 3. .....
U i i i I i I i i I
-9 -8 -7 -5 -5 -4 ~23 -2 -l 0 1
DC Voltage across Inverter

 

Figure 56 Shoot-Through Duty Cycle vs. DC Voltage Across Inverter

3.3.1 High voltage & Low voltage battery options.
As in section 3.2, both high voltage and low voltage battery options will work with this

topology. The low voltage option will require a DC/DC converter.

3.4 Input Rectifier/VSI to Output CSI with Z-Source impedance network

(battery in parallel with Input Converter)

This circuit architecture is re-shown here for convenience.

89

Source Inverter Traction Inverter

A A A f‘o’p - A
Traction
Generator! Jigfi JIJES Jg} E] l BL LIL 5L Motor/Re-

 

 

 

Starter Motor generative
+ x __
q, VI "
VG A Vrn

 

 

 

 

 

 

 

 

 

 

 

I? It I? m. at a a

Figure 57 Input VSI/Rectifier to Output CSI topology
Battery is parallel to the Input VSI

This topology is similar to the topology of section 3.2 with a few exceptions. In this
case, the battery is parallel to the input inverter rather than parallel to the Z-source
capacitors. The “Battery and Traction Motor mode” functions very much like that of
section 3.2. Also, the “Battery, Starter Motor and Traction Motor mode”, “Generator,
Battery Discharge and Traction Motor mode”, and “Generator, Battery Charging and
Traction Motor mode” each function very much like section 3.2. It is the “Regenerative
Braking and Battery Charging mode” that is different. During re-generative braking
mode, as before, the output inverter is controlled such that the stator field now resists the
rotation of the rotor and the voltage polarity across the traction inverter is now reversed
(positive terminal is down). The current in inductors L1 and L2 remains constant, and
the re-generative braking current flows through the Z-source capacitors and to the
battery. This occurs during the Active State of the Re-Generative Braking Mode.
Following this state, are the Open-Circuit Zero state and the Short-Circuit Zero state.
The duration of each state is controlled to achieve the desired level of braking resistance

and or the rate of battery charge. Desired braking resistance in excess of that available

90

via regenerative braking must be accomplished with the traditional mechanical friction
brakes. The mechanical fiiction brakes must be made an integral part of the hybrid
(friction & electro-mechanical) braking control system. A schematic of each of the three

regenerative braking states are shown below in Fig.58, 59, & 60.

Output CSI in Active:
Re-Generative Braking State

 

 

 

 

 

 

 

 

 

 

 

 

 

 

 

 

 

‘_ Traction Inverter
i + L1 A A 4 Traction
T» L L L Motorl Re-
‘ ‘ ‘ generative
2 Brake
Sv;1: —— 4:01 c+2 JUL/WV
A M vm
i l I I)IL gr 1>lL
-I> @ f v '

 

<——

.L2+

Figure 58 Regenerative Braking with VSI-Z—CSI and Battery Parallel to VSI Active
State

91

Output CSI in Open-Circuit Zero:
Re-Generative Braking State

 

 

 

 

 

 

 

 

 

 

 

 

 

 

 

 

 

 

_, Traction Inverter
- "66‘ - - - -
i + L1 - I I I I
I Re—generative
Brake
+ _l + +
ISVBf— for c2 GUI/Wi-
- "’ x a:
Vi crime—4%
M Vm
I t l
- rm - i i i
‘—
- 2 4-

Figure 59 Regenerative Braking with VSI-Z-CSI and Battery Parallel to VSI Open
Circuit Zero State

Output CSI in Short Circuit Zero:
Re-Generative Braking State

 

 

, Traction Inverter
A ’70:? Traction Motor!
i + U - Re-generative
I T Brake

 

W:-

W

 

 

 

4N
VI
<
W
A]
r I
\I
II
0
d
, O
N
\I
II

I
S +
\L AL
' II

\I

II

II

 

 

 

 

 

 

 

 

 

 

 

 

Figure 60 Regenerative Braking with VSI-Z-CSI and Battery Parallel to VSI, Short
Circuit Zero State

92

3.4.1 High voltage
The high voltage battery option will also work well with this topology. The
battery, as is shown in the schematics above, is protected with a series inductor. This
allows the magnitude of currents through the battery to be controlled, and it maintains the

current flow to the battery during the non-active states of regenerative braking.

3.4.2 Low voltage battery with DC/DC converter
The battery’s DC/DC buck boost converter would be connected parallel to the

input inverter just as the high voltage battery would be. An example schematic is shown

in Fig. 61 below.

Output CSI in Active:

 

 

 

 

 

 

 

 

 

 

 

 

 

 

 

 

 

 

 

Re-Generative Braking State
Low Volt Battery Option
+ _. _. Traction Inverter
i :13? 4 ‘ ‘ Traction
T I Ll 9 BL Motor! Re-
JQ} ‘ L l L l generative
+ 2: -—J1 52;: wwv 3"“
+ i ’ ' . :1: _‘L [.1 RT
Va]? x \g _: W
- - Vm
JUL/4W»
L L L
* 9 FL 9

 

-L2+

Figure 61 Input VSI to Output CSI with Z-Source DC Link
Low Voltage Battery Option with DC/DC converter

93

3.5 Summary

Each of the circuits show:

1.) Traditional Input Rectifier/V SI to Output VSI (VSI-VSI)

2.) Input Rectifier/VSI to Output CSI with Z-Source impedance
network (VSI-Z-CSI) (battery in parallel with Z’s capacitor)

3.) Input Rectifier/CSI to Output VSI with Z-Source impedance
network (C SI-Z-VSI) (battery in parallel with Z’s capacitor)

4.) Input Rectifier/VSI to Output CSI with Z-Source impedance
network (VSI-Z-C SI) (battery in parallel with Input Converter)

Have unique features which have been discussed. Circuit 1.) has the limitation of
only being able to buck voltage levels when providing power from the DC Link to the
traction motor. Circuit 2.) has the ability to both buck and boost voltage levels and with
its CSI output, it provides a smooth voltage wave to the traction motor. Circuit 3.) also
has the ability to buck and boost voltage levels. Its input CSI will have smooth voltage
wave forms for the input generator. However, since the traction motor must handle a
larger amount of power, the smooth voltages provided by a CSI would be of greater
advantage at the output. On the other hand, an output inverter must be sized for larger
shoot—through currents and a CSI must be sized for all of the shoot-through current to go
through one phase leg of the inverter. The shoot-through currents of a VSI are shared by

all three phase legs. The specifics of topology designs are presented in chapter 4.

94

4. COMPARISON OF POWER ELECTRONICS
TOPOLOGIES

4.1Methodology for Comparison

Various methods are available for comparing one circuit topology to another. A
comparison method that lends itself well to power electronics circuits is the determination
of the Switching Device Power of each different circuit. This is the method that will be

used in this analysis and it is discussed below.

4.2 Description of Switching Device Power Comparison

Switching Device Power (SDP) is a calculated value. The SDP rating of a circuit
is related to the cost of a design which will be explained in a moment. The SDP of a
single switching device is the product of the Voltage across a device during its open state

and the Current through the device during its closed state.
SDP = V OPEN * I CLOSED

Notice that this is not the power dissipated by the device, nor the power flow. The SDP
is a relative bench mark. Both current capacity of a device and voltage blocking ability
contribute to the size and cost of the device. As such, SDP can be used as a type of cost
comparison bench mark. The SDP rating of a circuit takes the SDP of each switching
device and aggregates them to arrive at the total for the whole circuit. The total SDP of a

circuit can then be compared to the total SDP rating of the other circuits.

95

To summarize:
Several factors contribute to the total SDP rating of a circuit.
Voltage stress on a switching device.
Current stress on a switching device
The total number of switching devices in the circuit

More on the Voltage Rating of a Switching Device

 

A semiconductor switching device has a voltage breakdown rating which is the
maximum voltage that a device can withstand while continuing to block the flow
of current. When the device is in its intended open state (gate voltage is below
Vg-on and the p-n junction remains reverse biased) only a negligible reverse
saturation current is allowed to flow. But when the voltage across the device
becomes to large, the device will begin to allow current due to thermal instability,
tunneling, or avalanche multiplication which is not desirable and will destroy
IGBTs or any semiconductor device which is not designed for this specific type of
current. The designed voltage rating of a switching device is directly related to its
size and cost. Hence, a circuit that can do the same job as another circuit at lower
switch size and cost is a preferable circuit design. This is one of the reasons why
the voltage stress that a particular circuit imposes across a switching device is

determined. This value is included in the SDP rating of a circuit design.

As shown above, the current stress is also a part of the SDP rating. The current that a

semiconductor can conduct in the closed switch state is also related to the cost and size.

Therefore, we also include a Circuit’s required current flow in the SDP calculation.

96

More on the Current Rating of a Switching Device

 

The amount of current that can flow through a bipolar junction type device (such
as that used in an IGBT) is directly related to the available number of charge
carriers in its PN junctions, its diffusion coefficient, junction area, and base width.
It is inversely related to the diffusion length in the base region. As such, the
material used in the PN junction and its size affect the current capacity of the

device and are directly related to cost and size.

Finally, the total number of switching devices that a circuit requires is included.
Therefore, a circuit that can do the same job with fewer switching devices receives a
more favorable impact on its SDP rating. The next sections derive the SDP equations for
the four sub-circuits included in this study.

DC/DC — Boost - Buck Converter

Traditional VSI Inverter

Traditional ZSI Inverter

Current Fed ZSI Inverter

4.3 DC/DC Boost — Buck Converter
For the low voltage battery option, a boost-buck converter is required. This converter
typically connects a 360 VDC battery to the 780V DC Link. To transfer power from the
batteries to the DC Link, the battery voltage must be boosted to the level of the DC Link.
To transfer power from the DC Link to the Batteries, the DC Link voltage must be

bucked.

97

 

DC-DC with

 

 

 

 

 

 

 

Boost Battery Buck
Storage
'80" ———¢
(+)/gnaw / 0
81 (+)
(+) ___ sz \ 7.: Va DC-Link
VBAT-"_ (-)
(-)
o

 

 

 

 

Figure 62 DC/DC Boost-Buck Converter

The voltage stress on each of the two Boost—Buck converter switches is VDc the DC Link

voltage. Average current through the switching devices is derived as follows.

First note that
P0... — Pin

VDCIo : VBAT 'IBAT
Po

[BAT : V
BAT

 

The two switches 81 and S; are operated in a complimentary manner (When 81 is open,

82 is closed and vice versa).

I

52 is average current through switch 82 , and

t2 on is the amount of time that S; is closed during a switching period TS,

Because of the large inductor in this circuit it can be assumed that 13 AT is the same during

98

each switch state (S; closed or Sz closed).

 

 

— _IBAT°t2_on
152- T
S
- _IBAT(TS—t2_on)
151- T
S

This gives the following
SDPAV : 152 'VDC +151'VDC

_ [BAT °t2_0n 'VDC + [BAT(T_t2_0n I/Dc

 

 

 

T5 T5
_ [BAT 't2_0n 'VDC +IBAT 'TS 'VDC "'IBAT 't2_on 'VDC
TS
: [BAT 'VDC

P0
SDPAV = T/_ VDC

BA T

99

4.4 Traditional VSI with PWM
To derive the SDP equation for the traditional VSI inverter, we will refer to the circuit

below.

 

 

 

 

 

 

 

" IE IT} 432:
322 C11:
P M
Vd
, . a a a

 

 

 

Figure 63 Traditional VSI

The voltage across an inverter switch is the DC link voltage. For the circuit topologies in
this study, the required voltage at the traction motor and at the generator are both 480V

line to line, RMS. Therefore:

2J5 ' VLL
«fiM

Where: VLL is the RMS line to line voltage at the motor
M is the modulation index of the inverter (Simple PWM)
VDC is the DC Link voltage

VDC :

The PWM of the inverter is operated in such a way that the constant VDC is impressed

100

across the load in pulses that average out to a sinusoidal wave form. Because of this and

the inductance ofthe load, the output current is simply:
‘5 - cos<¢> - V“

Line

 

Where: P 3a) is the 3 phase power delivered to the Motor

ILine is the Motor’s RMS line current that flows through the
IGBT

Note that ILine is an RMS value. To get the average current that the IGBT conducts, we
integrate the current sinusoid over one half cycle and divide by this period.
IL sm(a)t)da)t

ine

o'——-.>~1

 

A
av ILine

N.
||
||

>4 IN

72'

A

Where I Line is the peak line current

2J§P3¢

2
1 = 2—=
“V JE-cos(¢)-VDCJ§-M fr:

8PM
3n - eos(¢) - VDC ~M

 

 

The average SDP of the traction inverter is

_ 16R,¢
av _ 7r-cos(¢)-M

 

(SDP)av = 6.VDC I
Where 6 is the number of switches used in the inverter

101

4.5 Traditional Z-Source Inverter [ the Voltage Fed Z—Source Inverter (V F-ZSI) ]

with PWM

 

*L'- If} E} M}

41hP
o +
)L

I-
K

_ XCZ + ‘m’m
/ ' Vd ”(RPM—Ea)

 

 

 

 

 

 

 

 

 

t —v .

+L2-

Figure 64 Traditional Z-Source Inverter

SDPAV for the ZSI is based upon average current through the switching devices
times the voltage across the switching devices when they are open. This is the same as
with the traditional VSI topology. However, the ZSI has a few differences. First the ZSI
has all of the traditional switching states as the VSI, and it also has a short circuit zero
switching state. During this short circuit zero (SCZ) state, the current through the
switching device is ZILZ. During this SCZ switching state, all six switching devices are

gated on so that the current will be shared by all three phase legs of the inverter. As such,

102

l

the current in a switching device is ‘3‘ (211,2 ) and the average time period of this state is

Tscz/Ts-

Where: Ts is the switching period.
Tscz is the SCZ period.

Therefore the average switching device current contributed by the SCZ state is

2 I TSCZ _ _

3 L2 T . The other sw1tch states of the Z-source inverter are the same as the
S

traditional VSI (open circuit and active states). The current of these two states follow the

same equation as that of the VSI except that this current is averaged over a smaller

T S — T SCZ
portion of time which is the switching period minus the SCZ period TS

The average current is the sum of the current during these two time periods (SCZ time

period Tscz, and active + open time period (TS-Tgcz».

I =_2_ TSCZ 4P3¢ TS-TSCZ
AV

L2
3 T S 3VDCM7r 003(6) T 5
Where P3¢ is the power of the motor
M is the modulation index of the inverter (Vref/Vm)

I AV is average current

 

 

ILZ is inductor current
TSCZ is short circuit zero time period
TS is switching period

VDC is the DC link voltage

103

Due to the boost capability of the Z-source topology, the voltage across the switches is

not the DC input voltage. In this case the switch voltage is the boost factor times the

 

 

input DC Voltage
VDC : B ° VIN
Where:
VDC is the DC link voltage (the voltage across the inverter)
B is the boost factor, and
VIN is the input DC voltage
B = TS
_ [16] Therefore,
T S 2 T SCZ
T
_ S
VDC _ T 2T VI’V
S - SCZ
Where:

VDC is the voltage across the inverter.

Therefore, the average SDP of the Traction Motor Inverter of the Z-source topology is

2] Tsc‘z 4P3¢ TS—TSCZ TS
LZ

SDPAV = 6 — + Vm
3 T S 3VDCM7I cos(6) T S T S —2TSCZ

 

 

 

104

4.6 Current Fed Z-Source Inverter (CF-ZSI).

- "6‘0" - - -
+ L1 -
Sap BL prEIL 5“ng

 

 

 

 

 

 

\l
I

S +
II
)I

III
I

 

 

 

 

 

 

 

 

 

San I)“. Sbn Ph- Scn Pil-

Figure 65 Current Fed ZSI

 

 

This topology is very similar to the traditional ZSI. The difference is that the switching
devices of the CF-ZSI block reverse current flow. The traditional VSI-ZSI allows reverse

current flow due to the anti-parallel diode.

105

Antiparallel
Diode

Figure 66 IGBT Module Schematic

The switching device of the current fed ZSI does not allow reverse current flow and can
be modeled as shown in figure 67. This device is also called a non-punch through IGBT

(NPT-IGBT). [22]

Model of
I_Reverse

Blocking
IGBT

Figure 67 Reverse Blocking IGBT Module Schematic (RB-IGBT)

For this thesis, the following schematic symbol, as shown in figure 68, will be used to

represent this device.

Reverse
L Blocking
IGBT

Figure 68 RB-IGBT Symbol

Given a current source input, an inverter constructed of the reverse blocking IGBTS, will

function like a current source inverter. The CF-ZSI however, has the ability to both buck

and boost current (boost and buck voltage, respectively) where as the traditional CSI can

106

only buck current (boost voltage). For this discussion, the switching methodology for the
CSI is based upon a duality of the traditional Simple PWM switching for voltage source
inverters. The following duality will be used:
The list below shows traditional switching states of a VSI on the left and the
corresponding (dual) switch state used for the CSI on the right. Figure 69 shows

an example ofthe 01 l, 100 switching state.

 

\l
[I

 

W

Vd W

, W... VTM

 

 

0

 

 

 

- 1 ti“

San Sbn Scn

 

 

Figure 69 Illustration of the 011,100 switching vector

107

VSI CSI

OCZ states: SCZ states:
000, 111 orlll,000 :> 100,1000r010,0100r001,001

Active states:

100,011 100,001
110,001 010,001
010,101 010,100
011,100 001,100
001,110 001,010
101,010 100,010

Although the switching device and switch control methodologies are very similar to the
traditional IGBT, and traditional PWM used in V818 and VF-ZSIs; the equation for
current through the switch device is significantly different than that of the VF-ZSI. The
shoot-through mode (short circuit zero state (SCZ) ) of the previously discussed VF-ZSI
occurs during the open circuit zero (OCZ) states of operation. But the SCZ of the CF-ZSI
occurs during its active states. First the OCZ states and SCZ states of the VF-ZSI are

explained.

OPEN CIRCUIT ZERO, & SHORT CIRCUIT ZERO STATES OF THE VF-ZSI
These OCZ states occur when the positive half of the reference voltages (Va_ref, Vb_ref,
and Vc_ret) are less than the triangular carrier wave and when the negative half of the

reference voltages are greater than the triangular carrier wave. The SCZ states occur

108

when the PWM carrier wave is greater than the positive DC reference voltage and when

the PWM carrier wave is less than the negative DC reference voltage. See figure 69

Short
Sap 5 g Circuit
pr 5 l ' ' Zero
SCP . 5 Region
San '

Sbn
Scn

100.0
50.0
S 0.0
-50.0 Vb_ref
-l00.0 Vdc-H

 

 

I I I I I
49.8m 49.85m 49.9m 49.95m 50.0m “3)

Figure 70 SCZ Switch States of VF-ZSI

Sap through Scn represent the switch states of the inverter. See fig (69)
V_tri is the triangular reference voltage (PWM carrier wave)
Va_ref, Vb_ref, Vc_ref are sinusoidal switch state control reference voltages
Vdc_(+) and Vdc_(-) are DC reference voltages that control the SCZ switch states

109

Secondly, the SCZ states of the CF-ZSI are explained.

SHORT CIRCUIT ZERO STATES OF THE CF-ZSI

The CF-ZSI however has a different SCZ state. The SCZ of the CF-ZSI occurs during
the other wise active states. As before, the active states occur when the positive reference
voltages are greater than the triangular carrier wave and when the negative reference
voltages are less than the PWM carrier wave. However, the OCZ state now occurs when
Vdc_(+) reference voltage passes below the carrier wave and when Vdc_(-) passes above
the carrier wave. This is where the SCZ state had existed for the VF-ZSI inverter.
Further, the period of the SCZ varies during the cycle of the reference voltage. Also, all
of the SCZ current is conducted by each of the three phase legs of the inverter in turn,
rather than by all phase legs at the same time. First, phase “A” leg conducts the SCZ
current during the 30° to 90° portion of its reference voltage cycle, then the phase “B” leg
carries it during the 30° to 90° potion of its cycle, then in a similar manner, the phase “C”
leg will conduct. (With the VF -ZSI, all three phase legs conduct 1/3 of the SCZ current
during each carrier wave cycle). In the diagram below, it is the phase “A” leg that is
conducting the SCZ currents. As the phase “A” reference voltage moves through its
cycle from 30° to 90° the SCZ period is controlled by the moment that Vdc_+ and Va_ref

intersect with the positive portion of the carrier wave, and when Vb_ref and Vdc_(-)
intersect the negative part of the carrier wave. As Va_ref [Va SlII(Ct)t)J and as

A . 0
Vb_ref IV]; Sln(a)t _ 120 )J move through their cycles, the period of the SCZ

pulse changes.

110

 

Short Circuit Zero
Region

 

Sap
pr

Scp

San
Sbn
Sen

100.
50.
V_tri

E 0, Vc_ref

-50. Vb_ref

Vdc_(-)

 

-100.

 

I l l .
59.4m 59.45m 595m t(s)

Figure 71 SCZ Regions of the CF-ZSI

Sap through Sen represent the switch states of the inverter. See fig (70)
V_tri is the triangular reference voltage (PWM carrier wave)
Va_ref, Vb_ref, Vc_ref are sinusoidal switch state control reference voltages
Vdc_(+) and Vdc_(-) are DC reference voltages that control the SCZ switch states

In order to calculate the average current during SCZ, the total SCZ time summed over the

111

reference voltage period must first be determined and then divided by the period of the
reference voltage. This gives us the portion of the reference voltage wave form that is

used for the SC Z state. Since the SC Z current of the ZSI is twice the inductor current, the

- I = 2 * I * T
average SCZ current 18 as follows: SCZ _ Ave LZ SCZ _ AVG

Where
_k_ final r T "x.
T =< z: rel—SW I
SCZ .
k=o \ "I" ,/‘

 

./ I}: ref — lief sin(k(T3wa)m0m )+ %)\
“ +

r

./
2‘ _grl {1 dCJvf + IIrefSH (1‘ (Tmnmoror l+ 7% — 2%)
.-1

. . 1’
\~ '
/'\ I / -

a)
_ motor :1:
TSCZ_AVG ‘( ) TSCZ

 

 

 

272'

and

k = 0,1,2,..., k find,

_(%-%I

1 _
fi’” (Tm

 

motor )

112

Example:

lfa)

motor

_ _ rad
—1000rpm .. 104.72 A

And Ts = 0.0001

Then: kfinaF-‘IOO

11.1 = the impedance network inductor current

wmotor = electrical angular velocity of the traction motor or generator

fPWM = frequency of the carrier wave

Tscz = total short circuit zero time period during one cycle of the traction
motors voltage or current sine wave.

T5 = is the switching time period also know as the period of the PWM

carrier wave.

TSCZ_AVG = the proportion of time spent in the SCZ state.

Tscz_Avo*Ts = the average amount of SCZ time during one PWM
switching cycle.

VDC = the DC link voltage

A

ref = the peak voltage level of the PWM reference voltage which is

compared to the carrier wave to determine the required switch

state.

113

Notice
8181) of

TSCZ

increrr

%:

Vt = the peak voltage level of the PWM carrier wave.
Notice that two SC Z states occur for each half cycle of the carrier wave, and that each

step of k adds the time period of another one of these SCZ states.

1
T SCZ is a summation of each SCZ state from "/6 to ”/2 in A Tswwmotor sized

increments. Notice in fig. 71 that the period of time from point (1.) to point (2.) is

1
/2 TISW wmotor .

114

(V)

100.0 (1)
50.0
V_tri
0.0 Vc_ref
-501) Vb_ref
Vdc_(-) (
-100.0
I I F
59.4m 59.45m 59.5m t(s)

IActive_ Ave =

 

Short Circuit Zerol

Regon

//

    
  
  
  
 
   

Sap
pr
Scp
San

Sbn
Sen

 

Figure 72 SCZ Switch States of CF-ZSI detail

 

3VDCM7r 003(6)

810,,

 

1— T
3VDCM7r cos(6) ( SCZ—AVG)

115

Ts

8P3¢ [Ts _TSCZ_AVGTS

J-

And the total average current through the switching devices is the weighted sum of the

(1)active and open state currents with (2)the SCZ currents.

Iave_SCZ 2 2[L2 Tscz _AVG

810,,

I +
3VDCM75 cos(6)

 

Ave

=21L2Tscz_.4VG (1_TSCZ_AVG)

V

Where DC : Vc

f/‘C_

_ The average voltage across the Z network capacitor.

The equation for voltage across the switching devices is similar to that of the Voltage Fed

ZSI. Tscz OfIhC VSI ZSI IS replaced by TSCZ_AVG*TS

Ts Ts
VIN = ,,.
Ts-ZTSCZ_AVG Ts

 

 

VDC = VIN
Ts - 2TSCZ

Where: VIN is the input voltage

VS = VDC is the voltage across the switching devices.

As a result, the SDP of the Current Fed ZSI which has 6 switches is

SDPCF—ZSI = 6 * VS * I A VE

V 2 8P,
SDP - =6 W —1 T + t l—T ,
(“F-ZSI [1_ 2TSCZ_AVG ][ 3 L2 SCZ_AIG 3VDCM7I COS(6)( SCZ_AIG )j

 

 

116

4.7Applieation of Equations to the Back to Back Topologies
Maximum voltage and average currents have been calculated for the various sub-circuits
included in this study. These can now be combined to form the SDP equations of the

complete back-to-back inverter circuits.

4.7.1 Back to Back VSI-to-VSI Network Topology

Traction
Motor! Re-
m Brake
LT RT
dam—3(3)
Vm
480 V
-100 to
+250 kW

 

generative

- 41"}

fl

 

S_3T-p
SéT-n

 

 

st = 10 kHz

 

81 T41 SZr-n

 

Traction Inverter
SjT-p SZT-p

1:}
a}

Vd

 

Cs

 

 

 

 

 

Source Inverter
sas-p She-p SCs-p
.16} J@ J@

I):
sas-n She-n SCs-n

Few = 5 kHz

 

 

Imp

   
 

Generator
I Starter
360 to 480 V
0 8. 120 Hz
0 to 150 kw

Figure 73 Back to Back VSI-to-VSI Network Topology

117

The Back
generator
inverters

Since the
in the eq
motor‘ger
motor ’ ger

variables .

The Back to Back VSI-to-VSI Network Topology shown above uses a VSI inverter at the
generator end and at the traction motor end of the circuit. Each of these traditional VSI
inverters function in the same way and the SDP is calculated in the same way for both.
Since the power levels are different from one end to the other, different variables are used
in the equations. For the following discussion, it will be assumed that the 3 phase
motor/generator at the left end is the generator connected to the diesel engine and the
motor/generator (motor/brake) at the right end is the traction motor. The following

variables are hereby added or clarified:

MG is the modulation index of the generator inverter
MTM is the modulation index of the traction motor inverter

P 3¢-G is the maximum 3 phase power delivered by the
Generator

P3¢-TM is the maximum 3 phase power delivered to
the Traction Motor

IL_G is the RMS Generator line current that flows through
the generator inverter IGBTs

IL_TM is the RMS Traction Motor line current that flows
through the traction motor inverter IGBTs.

[avg is the average current through the switching devices of
the generator inverter

IaV-TM is the average current through the switching devices
of the traction motor

(SDP)aV-G Is the SDP rating of the generator inverter

(SDP)aV_TM Is the SDP rating of the traction motor inverter

118

(SDP)aV-Dc DC Is the SDP rating ofthe DC/DC converter

Therefore,

16F..-
”—0 _ nocos(¢)-MG

 

(SDP)av—G : 6 VDC 01

The combination of a VSI at both ends of the voltage source DC link is just a
matter of summing the individual equations as follows (traction motor inverter, generator

inverter, and DC /DC converter).

SDP A V—VSI—VSI :
(SDP)av—Trl/I + (SDP)av—G + (SDP)av—DC/DC :

161004,, + 16103.54 + 11.3.“
fl - COS(¢) ° Mm 71' ’ COS(¢) ' MG VBAT

 

 

DC

4.7.2 Back to Back VSI-to—CSI Z-Network Topology (V SI-Z-CSI)
In the following topology, a traditional VSI inverter, which uses the standard free
wheeling diode anti-paralleled with an IGBT, is used at the generator end of the circuit.
A reverse blocking IGBT is used in the inverter at the traction motor end of the circuit.
The impedance network (L1, Cl, L2, C2) is between the traditional inverter and the
reverse blocking inverter. We will interchangeably call this the “Back to Back VSI-to-

CSI Z Network Topology” or “VSI-Z-C SI”.

119

 

 

0
83-5.» >5 5
5‘83 o
0308 >‘°
Eva-cm N
i—Oaa 0+
EC» 3.8
in? 8
I.—_1(
o——{(—o——Ie——0
D

 

 

 

 

 

 

 

Traction Inverter
S1r-p $211; S3r-p

v

 

 

sm sin sari
st= 10 kHz

 

v

 

 

‘01 cr-
”611‘
- L2 +

 

J

 

'

It

 

 

 

'

 

Source Inverter
Sis-p 505p SCs-p
Sam She-n SCs-n

 

 

st= 5 kHz

E

Figure 74 VSI-Z-CSI

 

 

 

 

(D
>

Generator
IStarter Wt as

0&120Hz
0to150kW

Motor Rs Ls
@

36010 480V

The traditional inverter connected to the generator is primarily operated as a controlled
rectifier which delivers the generator’s power to the rest of the circuit. This controlled
rectifier’s switching states must be coordinated with those of the traction motor inverter,

however. When the traction motor inverter goes into its open circuit state, 21L will flow

120

through the freewheeling diodes of the generator inverter.

 

 

 

 

 

 

 

 

 

 

 

 

 

 

 

 

 

 

 

 

 

 

 

 

 

 

 

 

 

 

 

 

 

 

 

 

+ +
RS LS +—_I——;LC1 csz + LT RT
|W’ as VBI— ' J V32 ;: L
[AL b5 - - ;:
VG Vm
cs
e e a
fi ' v fl - L2 +
Figure 75 VSI-Z-CSI; CSI in OCZ State
- - ~ rim
‘3; ’3} £2: + L1
.1 .l .1
Rs Ls 5.1+ +172.— * LT R
MID—0 as VB1T_AC1 CI‘_T V82 7:
(Av; W” “5 b5 - - :: M
VG Vm
' 7 L2 T
Figure 76 VSI-Z-CSI; CSI in SCZ State
[AV G 52.111 TOCZ—TM + 8PMS-G Ts "TOCZ—TM
— 3 TS 3VDC_GMGfl'COS(6) TS

Where:

IAV is average current through the switching devices
1L2 is inductor current of the Z network
Tocz-TM is open circuit zero time period of the traction motor

inverter which is also the SC Z time period of the generator

inverter.

121

TS is switching period which is the same for both inverters

P3¢-G is the three phase power being generated by the generator
MO is the modulation index of the generator inverter

Vooo is the DC voltage across the generator inverter

V _ VLH, 272
And in this case, DC-G _ Therefore,
75M,

 

[AV 0:31” TOCZ—TM + ”RPM-G Ts—TOCZ—TM
_ 3 TS 372' COS(9)VLL_G TS

The Traction Motor Voltage is:
Ts
Ts — 2TSCZ-TM

 

l7DC—TM = 7C = B ' VDC-G = VDC—G

Where
Tscz-TM = SCZ time period of the traction motor inverter.

_ Ts
Ts TZTSCZ—TM

 

and as shown above,

V = VLH; 272
DC—G ‘/—
3M6

 

122

SDP of the generator inverter is

 

T ,, HEP _ T —T
SDPav—G =6 3,142 OCZ—TM + 3415 G s OCZ—TM *
3 TS 371' COS(6)VLL_G TS

[ Ts VLL—G 2‘5]
Ts ‘2TSCZ_TM fiMG

And the SDP for the current fed ZSI has already been discussed and is shown here

with the appropriate variables for the VSI-Z-CSI Network circuit topology.

SDPAV—TM = 6 * VSW * [AVE : 6Vd—TM *

_2_ [L z TSCZ-TM + 8P3¢-TM 1_ T SCZ-TM
3 TS 3VDC—TMMTM” 008(9) TS

 

 

 

For the case where one DC/DC converter is used and it is in parallel with
just one of the Z network capacitors, the SDP of the DC/DC converter was also shown

before and is

Po- P..-
SDPAV—DC/DC = in : AB ' VDC—G

 

 

 

VBAT VBAT
V BAT Ts — 2TSCZ-TM J3MG

The SDP of the combined topology is

SDPAV—VSI-Z—CSI = SDPAV—G + SDPAV—TM + SDPAV—DC/DC

123

SDPAV—VSI—Z—CSI =

6 3 [L2 TOCZJM + 2J9P3¢—G Ts ‘TOCZ—TM 2~/§VLL—(; +
3 TS 3n cos(6)VLL_G TS fiMG

6K2‘EVLL—G I Ts j}:
JEMG TS_2TSCZ

31 Tscz-TM + 8P3¢—TM 1_ T SCZ—TM +
LZ
3 TS 3VDC—TMMTM7ICOS(6) TS

Po—BAT ( TS )VLL—Gz‘fi
VBAT Ts -2TSCZ 5M0

 

 

 

 

 

124

 

4.7.3 Back to Back CSI-to-VSI Z-Network Topology

generative
9
Vm

Traction
Motor I Re-

 

 

480V

Traction Inverter
SITp 321p 8311:
Q #1; «T3 -100to +250 kw
SIT n SIT-n S3T-n
st=10kllz

 

 

M Brake
l-r Rr
tl—kllll/IW

v

 

ti

 

 

 

v

 

Ii

 

 

 

 

L2+

 

 

 

 

 

 

 

 

- #4 1i“ E

E ‘3... , ‘2’

3 £154 3" 53> a
”Hit-4 0—l(——0—I ¥ "’5

 

 

 

 

 

——-l( {—0
t: E

>

36010480V
0&120Hz
0101ka

Figure 77 CSI-Z-VSI

This arrangement places the inverter with the reverse blocking IGBTs (CSI) at the
generator end of the circuit. Therefore, the CSI functions primarily as a controlled

rectifier. As before, the generators controlled rectifier must be operated in a way that

125

coordinates switching states with the VSI at the traction motor end of the circuit. During
the traction motors open circuit zero states, the generator’s inverter must be in a SCZ
state or an active state. It is during this state that the generator CSI switching devices
conduct ZILZ. Unlike the traditional VSI type of inverter with switching modules that do
not block reverse current, only one phase leg can be operated in the SCZ state at a time.
Therefore one phase leg must handle the entire 21Lz current. However, the phase leg that
is used for this is rotated from one SC Z state to the next. As a result, the average SCZ
current that a phase leg must conduct is still 1/3 of the 21Lz- Also recall that the SCZ
current flowing through the generator’s switching devices flow only during the traction

motor’s open circuit zero state ( TOCZ-TM ).

2 T
IA V—SCZ—G : _ ILZ OCZ—TM
3 T 5

Where 1 AV is average current through the switching devices
In is inductor current of the Z network
Toczm is open circuit zero time period of the traction motor
inverter which causes 21L; to flow through the generator switch

devices

TS is switching period which is the same for both inverters

The active state switching device current of the generator’s CSI is

= 4P3¢-G Ts - TOCZ—TM
3Vd—GMG” cos(t9) TS

 

IA V—Active—G

Where: Vd-o is the DC voltage across the generator inverter.

126

 

 

 

Val—G
fiMG
I . : 4fiMGP3¢—G Ts - T OCZ—TM
A V—Actli‘e—G 3 . ZEVLL—GMGflCOS(6) TS

 

I _ TERM-G [Ts _TOCZ—TM]
A V—Active—G _

3VLL_G7Z'COS(6) TS

Therefore, the average current for active and SC Z states are:

I _ = —I
AVG LZ TS

2 T1—TM + HEP-3W0 Ts ‘TOCZ—TM
3 TS 3VLL-G7’ 008(9)

The average current for the traction motor VSI and for the DC/DC converter have

been derived before and the voltage across the generator inverter switches is

2V 2VLL—G 2V
JEMG C'

These are:

 

 

 

 

_ 2 TSCZ’TM 8P3¢-TM Ts - T0—TM
1A V—TM — —1 L +
3 T S 3VdCMTM7rcos(6) T S

Ts
Ts "ZTSCZ—TM

 

Where: Vdc—TM : VC : VdCTG

127

 

I _ PO—BAT
AV—DC/DC — V
BAT

V . = 2‘/§'VLL—G TS
dC—TM 6M G Ts — 2TSCZ—TM

 

Therefore, the SDP of the back to back C SI-Z-VSI topology is

SDP A V—CSI—Z—VSI : SDPav—G + SDPav—TM + SDP A V—DC/DC

SDP A V—CSI—Z—VSI :

 

 

 

6 2 ’ILZ TOCZ— TM+ NEPM—G Ts "'TOCZ—TM a:
3 +3VLL_G7Z'COS(6) TS
|:2\/—23—::4LLG— —G ]:
6 31L ZTSCZ— TM+ 8P3¢-TM Ts — TSCZ—TM *
3 +3Vdc—TMMTM7ICOS(6) TS

[ Ts 2‘/—VLL— S—JFG]

Ts 2Tscz [MG

[Po-BAT Z‘fz—VLL—G( TS H

VBAT fiMG TS _ZTSCZ

128

4.7.4 Comparison of Circuit Topologies
To help facilitate a comparison, the SDP of each topology is restated here.

Back to Back VSI-to—VSI Network Topology

(SDP)av—TM + (SDP)av—G + (SDP)av—DC/DC :

 

16P3¢_TM 16P3¢—G PO—DC/DC
+ + VDC
7r - COS(¢) ° M TM 7! ' COS(¢) 'M G VBAT

Back to Back VSI-to-CSI Z-Network Topology

SDPAV—I'Sl—Z-CSI :

 

6 3 1L2 TOCZ_TM + ZN/ng—G Ts ‘TOCZ—TM 2«El/Lao +
3 TS 37ICOS(6)VLL_G TS fiMG

6[[ 2~/§VLL—G I T s ]] ...
fiMG Ts — 2TSCZ-TM

31 TSCZ—TM + 8P3¢-TM 1_ TSCZ—TM +
LZ ___“ _—
3 TS 3VDC-TMMTM” COS(9) TS

P0—BAT( Ts JVLL—GZ‘E
VBAT Ts ‘2TSCZ—TM \[3—MG

 

 

 

 

129

Back to Back CSI-to-VSI Z-Network Topology

SDPAV—CSI—z—m Z

 

 

 

6 3] TOCZ‘TM + 2J6P3¢_G T S —TOCZ—TM zfiVLL—G +
3 L2 T5 3VLL_Grrcos(6) TS 6M0

6 3le TSCZ—TM + 8P3¢-TM Ts ‘TSCZ—TM
3 T5 3V dc—TMM TM” C08(9) T 5

 

Ts *2TSCZ—TM 6M0

 

[ Ts zfiVLL—G]+

PO—BAT 2fiVLL-G{ Ts H

VBAT fiMG TS-ZTSCZ—TM

Comparison
It is now desired to compare each of these SDP equations. To do so, a basic
operating scenario is considered. A typical set of operating parameters is the

following:

If we ignore the traction motor inverter losses, than power entering this

inverter is equal to the power leaving the inverter. Therefore,

P DC Z 3¢~TM
VDC—TMI DC Z P3¢—TM
r _ P3.”
DC '-
DC-TM

130

 

Where [DC is the average DC current entering the traction motor

inverter.

In steady state, the average Z network capacitor current is equal to zero. Since the
positive rail of the inverter is connected to the Z network inductor, capacitor
junction, the average current there is the sum of the average inductor current plus

the average capacitor current.

 

 

 

 

 

 

 

 

 

 

 

 

- Wk 1 ~
L1 ' /l\ i 1
II ;
+ + a C2 l i
1 c l Inverter ‘_ can.»
+ I "3
Vi I f:
— ' i
I
.. - t .
- L2 + 'L I
Figure 78 ILZ Average

Since 1c average = 0, then ILz average = Ioc_mv average. The Z network inductor

current can not change instantaneously and is close to a constant over a switching

cycle. ILZ can be approximated as ILz_an.

Also note that the inductor of the Z network delivers current to the traction motor

only during the active state ( not during the OCZ or SCZ states). If we let

131

 

1 DC _ INV be the average current in the DC rail that is being delivered to the

traction motor inverter, than

1 TACT _ 1 TS “TSCZ—-AVG ‘TOCZ—TM _ 1-
LZ _ avg T _ LZ _ avg _ DC INV
s _

 

Ts

Rearranging we can show,

 

1 = 1' TS
LZ_avg DC_1NV T T T -
S _ SCZ—AVG _ OCZ—TM

So that we can determine [M m, we now seek to findIDC m, in terms of the

traction motor power.
P DC :PAC

:1:
VDC_INV IDC_I..W P3¢—TM

VLL—TM 2J5 *1 U
\[3—M DC_IM

 

P3¢—TM

PW 6M
VLL—TM 2J5

 

I DC_1NV Z

Restating the equation from above,

 

_. Ts
ILZ_avg =IDC_1NV T T T
s — SCZ—AVG — OCZ—TM

And substituting,

P3¢—TM 43M [ TS )

VLL-TM 2J5 Ts “TSCZ—AVG _TOCZ—TM

 

 

ILZ _at‘g Z

132

Now we have all of the equations that we need to prepare a comparison by

using the basic operating condition shown just below.

P base = 200,000.
v base (L-L) = 480
v base DC = 480

1 base = 240.5

The following typical operating conditions are also defined.

MG 1. pu
MTM 1. pu
P3CD-TM 1. pu
P3q)_G 0.5 pu
PO-BAT = PDC/DC 0.5 pu
TS 0.0001 5
VB AT 0.75 pu
VLL-G 0.5 pu
VLL-TM 0.5 pu
wmotor 314. rad/s
o 0.451 rad

Using these values, the following SDP evaluations are determined.
SDPA V—VSI—VSI = 9.0
SDPAV—VSI—Z—CSI = 9-0

SDPAV—CSI—Z—VSI = 9.0

This shows the direct comparability of the Z network topologies against the

133

traditional Voltage Source topology. Yet the Z network topologies offer the
ability to boost or buck voltage levels from one end of the DC link to the other.
This improves the available options for maintaining different voltage levels at the
generator and traction motor while independently controlling power flow to or
from the generator, traction motor, and battery. The VSI-VSI topology is not able
to allow different voltage levels at opposite ends of the DC link. The Z network
topologies also enjoy an improvement in reliability due to both OCZ and SCZ
switching states being allowed. To prevent destruction of the switching devices,
these states are forbidden by the current source network and voltage source
network, respectively. Below in fig. 78 it is shown that the SDP comparison of
the topologies is very similar even when the Z network employs its unique ability
to boost voltage levels. For fig. 78, the same parameters as above were observed,
except that the traction motor voltage (VLL-TM)was varied from 0.5 to 1.0 pu. That
is, the generator voltage level was held at 0.5 pu while the traction motor voltage

level was varied.

134

 

 

 

 

 

 

 

 

 

 

 

 

 

 

 

 

 

 

 

SDP (PU) VS. VLL_TM
120 .1 , , . . . , . --..-..-__..._...___ __,-_-,.--__...._,._-_........ __._. ..___ ._ _ -1!
11.5
11.0
A I
a 10.5 ”f.
v '1‘ +VSIV3|
3 1° ° fix-'3‘ +VS|~Z~CS|
to 9 5 mad-J. "Z -'* ‘ +CSIZ-VSI
g 0 -__l=£!— Z fr—
8.5
8.0 I I I I r 1 I T I I
0.5 0.5 0.6 0.0 0.5 0.? 0.7 0.8 0.8 0.9 1.0
Sweep V-LL_TM (PU)

 

Figure 79 SDP Comparison

Perhaps the similarity between the comparisons is even more obvious if we display SDP
on a normalized basis. Figure 79 shows this. From this graph, we see that the Z-Network
tOpologies have an SDP of 1.17 at full boost as compared to an SDP of 1.06 for the non-

Z-network topology -

135

SDP
Normalized

+VSIVSI
4p—VSLZCSI
+CSl-Z-VSI

0.5 [16 0.6 0.8 Us? 0.? 0.8 0.8 019 1.0

Sweep V-LL_TM (PU)

 

Figure 80 SDP (Normalized) Comparison

This change in voltage levels was accomplished by the Z-Network topologies while

holding the modulation index of both the Generator and Traction Motor inverters at one.

This is not possible with the VSI-VSI topology.

Maintaining the modulation index at one (100%) reduces voltage stress on the devices by
allowing the DC Link voltage level to be kept at a minimum level for the corresponding
output AC voltage levels. A modulation index of one also minimizes the harmonic losses

in the generator and traction motor.[20]

136

A list is provided below to summarize the Pros and Cons of the topologies.

SUMMARY LIST
VSI-VSI topology

Pros:

Cons:

Z-Source topologies

Pros:

Cons:

There is no need to coordinate the switch states of the
generator inverter with the traction motor inverter.
Non-reverse blocking switching devices can be used in
both the generator and traction motor inverters.

Generator Inverter can only boost toward the DC Link.
Generator Inverter can only buck toward the generator.

An accidental shoot through would be destructive to the
switching devices.

This topology has the ability to both buck and boost voltage
levels.

This topology has the ability of maintaining different
voltage levels at both ends of the topology. This provides
an extra degree of control freedom which can be used to
lower stress levels on the switching devices and reduce
harmonics in the AC machines.

There is no risk of shoot-through for the VSI which
improves the dependability of this topology.

There is no risk of open circuit for the CSI which improves
the dependability of this topology.

A short circuit zero state of one inverter needs to coincide
with an open circuit zero state of the other inverter, and
vice versa.

137

4.8 Future Work

Much has been learned through the works sited above, and this has contributed greatly to
this research. In the process of this work, it has been discovered that additional work is
needed regarding the loss effects of the combination of inverters and AC machines. The
work of others has shown much research about loss effects in power electronics, and
much about loss effects in AC machines. A significantly lower amount of work has been
done regarding the loss effects of the interaction of these two systems. The research that
has been done in this area shows that significant progress may be possible through further
research of these topics. Combining power electronics with AC machines has vastly
improved the amount of control we have over various systems and this has allowed us to
improve operations and conserve energy. However, the switching methods of the power
electronic inverters cause harmonics in the AC machines and therefore iron losses.
Changes in these switching methods, changes in the way that motors are controlled, and
changes in the way that voltage levels are controlled by the power electronics may offer

significant improvements over the presently existing technology.

It has been found that the harmonic content of the inverters are lowest at a modulation
index of 100% and it is proposed here that a reduction in losses can be achieved by

Operating the inverters, which are driving the traction motors, at a modulation index of

138

100%. Including a DC/ DC converter in the above topologies may reduce losses in the

AC machines. See Figure 80 below.

Generator
I Starter

__

M=1

_L_J
Source

Inverter

Buck
(--)
Boost
(n)

 

 

 

 

 

 

 

1% 1111?

Motor Em
VG 1m

 

 

o——————-1||}——————<-

 

 

DCIDC
Converter

 

M81

Traction
Inverter

 

 

 

 

 

 

 

 

 

 

 

 

 

 

Figure 81 Proposed Topology

Traction
Motor I Re-
generative

Brake
’L

Vm

By placing a DC/DC converter in the DC link, the traction motor voltage magnitude is

controlled by varying the DC link voltage level rather than by varying the modulation

index of the inverter. The modulation index is held constant at 100% even while

implementing constant volts/hertz operation of the traction motor. Several topologies

should be considered.

139

~ Buck -) r—j

M=1 Boost (- MT“,
Source Traction

Inverter # ‘ Inverter
~ 1’ ' Q .1 Traction
M} J‘ ‘ Motor/Re-
generative

ww ke

Wm;

i Wm
12: J

42:13;

    

 

Generator
l Starterw J

Motor '
”\4 ‘W-
VG

“M

 

 

 

 

 

 
  
  

Boost -) J
DC'DC Buck 6

Converter L
with Low VoltL LV-
Battery % Batt

Figure 82 A Possible Implementation of the Proposed Topology
Using Traditional Uni-Directional Buck/Boost Converters

‘ Buck 03
M=1‘ Boost 0a M=11
Source Traction

 
 
  

 

 

Traction
Generator/ Motor / Re-
Starter generative
Motor ' ‘ Brake
Vm

 

 

Figure 83 A Possible Implementation of the Proposed Topology
Using Traditional Bi-Directional Buck/Boost Converter

140

»———1 Buck 1
M=1: (--) M=1'

1
I

 

 

Sioirfice 30054 Trafiion
Inverter , (‘9 “J Alnvegter - T
. raction
Generator! J6; Jig} 4‘ J9} 113} 1.1%} Motor/Re-
Starter

 

 

generative

Motor m x x Brake
frrx

VG am 1 Vrn

$113”? 11% «a ,4,

Figure 84 A Possible Implementation of the Proposed Topology

 

 

 

 

 

 

 

 

 

 

 

 

 

 

 

Using a Bi—Directional Qusi-Z-Source
Buck/Boost Converter

141

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