A THZ FOCAL PLANE IMAGING ARRAY USING METAMATERIALINSPIRED BOLOMETER AND WAFER-LEVEL INTEGRATED FOCUSING
ELEMENTS
By
Kyoung Youl Park

A DISSERTATION
Submitted to
Michigan State University
in partial fulfillment of the requirements
for the degree of

Electrical Engineering ‒ Doctor of Philosophy
2014

ABSTRACT
A THZ FOCAL PLANE IMAGING ARRAY USING METAMATERIAL-INSPIRED
BOLOMETER AND WAFER-LEVEL INTEGRATED FOCUSING ELEMENTS
By
Kyoung Youl Park
There is substantial interest in terahertz (THz) for applications in communications,
sensing, spectroscopy, imaging, and security. Existing THz systems are built using quasi-optical
setups. To reduce cost and make THz integrated circuits a reality, approaches to wafer level
integration of components is critically needed. Among the many THz systems, imagers are most
desirable as they can be immediately adopted for applications in medical-imaging, security, and
non-destructive evaluation (NDE). Many types of detectors have been studied in the design and
fabrication of THz focal plane arrays such as Schottky diode rectifiers and bolometers. A typical
focal plane array (FPA) pixel element consists of a read-out circuitry, detector device and an
antenna or a lens element. A 100×100 element FPA will require 10,000 of these individual
elements. Pick-and-place of such a large number of elements within a small foot print is cost
prohibitive and technically challenging. Wafer level integration of these elements is desirable to
overcome this challenge.
Among the many detector elements, bolometers are desirable as they are simple to
implement and has the potential to provide high sensitivity. At THz frequencies a bolometer has
to be directly coupled to an antenna element to achieve high coupling efficiency. However,
antennas thermally load the bolometers and reduce the overall sensitivity. Thus, new approaches
to integrating bolometer with antenna elements are needed to achieve high sensitivity.
Furthermore, new approaches to improving intrinsic sensitivity of a bolometer in the THz
spectral region are needed.

Key focus of this research is towards the design and demonstration of THz metamaterial
based absorbing structures and their utilization in the design of absorbers for packaging and
bolometers for focal plane arrays (FPAs), with major research focus on the design and
implementation of THz focal plane arrays. A range of bolometer designs based on metamaterial
structures are investigated, including cross, circles, Minkowski, and slit ring resonator which are
implemented using high resistive thin metal films. Minituarized unitcells are designed, fabricated
and tested in order to improve the sensitivity of the bolometers.
For beam focusing and enhanced coupling efficiency to the detector elements
(bolometers), wide-band micro-lens array are designed and demonstrated. These microlens
arrays can be fabricated at the wafer level using 3D printing. Furthermore, a new plasmonic
antenna element is demonstrated that can also be utilized in place of the conventional micro-lens
design. This antenna element is integrated in close proximity to the bolometer structures while
avoiding thermal loading the detector element. Performance of this antenna element is presented
in comparison to conventional antenna elements. A THz imaging array that integrates the
detector elements and lens element at the wafer level is demonstrated as part of this research
work. Apart from imaging array, under this research work, approaches to fabricating 3D THz
components at the wafer level have been demonstrated through the use of 3D printing and deep
metal etching processes.

Copyright by
KYOUNG YOUL PARK
2014

To my wife, Hee Young and lovely son, Jun-Beom
and
Parents

v

ACKNOWLEDGEMENTS
It is time to finish the long journey to the doctoral degree. Many people have supported
and helped me in completing my dissertation, whether directly or indirectly. I would like to
express my heartfelt gratitude to them for their contributions.
First of all, I would like to thank my advisor, Dr. Premjeet Chahal, for his guidance,
and support. His understanding and passion has helped me overcome many research challenges
during my pursuit of Ph.D. Also, he has taught me so many interdisciplinary theories related to
materials, microfabrication, optics and electromagnetics to broaden my engineering knowledge
and vision.
I would like to thank my committee members Dr. Edward Rothwell, Dr. Tim Hogan,
and Dr. Jeff Sakamoto, for the time they invested in helping me complete this dissertation.
Additionally, I received help and guidance along the years from Dr. Shanker Barasubramanian. I
send many thanks to lab-mates past and present: Dr. Nophadon Wiwatcharagoses, Dr. Jose
Hejase, Xianbo Yang, Joshua Myers, Amanpreet Kaur, Ifwat Ghazali, Steve Zajac, Cecila
Acosta and Jennifer Byford. Also, I would like to thank members of EM and NDE research
groups. Also, special thanks to Brian Wright, Gregg Mulder, and Karl Dersch for their avail to
my teaching class, experiments and microfabrication.
Furthermore, I would like to express my thanks to the Agency for Defense
Development, Republic of Korea for providing me the financial and moral support for my Ph.D
studies in the United States.
And, finally my deepest gratitude goes out to my family, who has always supported me,
my wife, Hee Young and my great son, Jun-Beom.

vi

TABLE OF CONTENTS

LIST OF TABLES…………………………………………………………………………….x
LIST OF FIGURES…………………………………………………………………………..xi
CHAPTER 1. THZ IMAGING ARRAY TECHNOLOGY .................................1
1. 1. Introduction ......................................................................................................................... 1
1. 2. THz Imaging system ........................................................................................................... 3
1. 3. Components for THz imaging system ................................................................................ 6
1.3.1 THz detectors ................................................................................................................. 6
1.3.2 THz focal plane arrays ................................................................................................. 12
1. 4. Goals of the dissertation ................................................................................................... 14
1. 5. Overview of the dissertation ............................................................................................. 17

CHAPTER 2. PERIODIC STRUCTURES: FREQUENCY SELECTIVE AND
METAMATERIAL................................................................................................19
2. 1 Periodic structure ............................................................................................................. 19
2.1.1 Properties of periodic structure .................................................................................... 19
2.1.2 Analysis of periodic structure ...................................................................................... 21

CHAPTER 3. TERAHERTZ ABSORBERS ......................................................25
3. 1. Metamaterial structure ...................................................................................................... 25
3. 2. Skin Effect ........................................................................................................................ 27
3. 3. Designs and simulations ................................................................................................... 34
3.3.1 Single layer proposed metamaterial arrays .................................................................. 34
3.3.2 Double layers of proposed metamaterial arrays ........................................................... 43
3. 4. Fabrication ........................................................................................................................ 49
3. 5. Experimental measurement............................................................................................... 51
3. 6. Measured results ............................................................................................................... 54
3.6.1 Reflection and transmission of proposed absorbers with various incident angles....... 54
3.6.2 Waveguide measurement of proposed absorbers ......................................................... 58
3. 7. Conclusions ....................................................................................................................... 63

CHAPTER 4. THZ POWER METER.................................................................64
4. 1. Designs and simulations ................................................................................................... 64
4. 2. Fabrication ........................................................................................................................ 69
4. 3. Experimental setup and measurement results ................................................................... 70
4. 4. Conclusions ....................................................................................................................... 77

CHAPTER 5. TERAHERTZ SURFACE PLASMONICS BULLS-EYE
ANTENNA ..............................................................................................................78
vii

5. 1. Theoretical background .................................................................................................... 78
5. 2. Metallic Bulls-eye antenna ............................................................................................... 80
5.2.1 Design and simulation.................................................................................................. 80
5.2.2 Fabrication ................................................................................................................... 87
5.2.2.1 Layout of the surface plasmon-assisted THz imaging array…………………...87
5.2.2.2 Metamaterial-inspired bolometer………………………….…………………...88
5.2.2.3 Fabrication of the bull-eye plasmon-assisted antenna…….…………………...89
5.2.2.4 Fabrication of the metamateiral-inspired bolometer….…….……………........89
5.2.3 Experimental setup....................................................................................................... 90
5.2.4 Measured results .......................................................................................................... 93
5.2.4.1 Characterization of the bulls-eye plasmonic antenna……..……………………93
5.2.4.2 Measurement results of the proposed 100 GHz imaging array..………………96
5. 3. Plastic Bulls-eye antenna .................................................................................................. 98
5.3.1 Design and simulation.................................................................................................. 98
5.3.2 Fabrication ................................................................................................................. 100
5.3.3 Measurement setup .................................................................................................... 101
5.3.4 Measured results ........................................................................................................ 102
5. 4. Conclusions ..................................................................................................................... 105

CHAPTER 6. DIELECTRIC MICRO LENS ARRAY FOR THZ FOCAL
PLANE ..................................................................................................................106
6. 1. Theory of Lens ................................................................................................................ 106
6. 2. Simulation ....................................................................................................................... 108
6. 3. Fabrication ...................................................................................................................... 117
6. 4. Experimental measurement setups.................................................................................. 119
6. 5. Measurement results ....................................................................................................... 122
6. 6. Conclusions ..................................................................................................................... 128

CHAPTER 7. THZ IMAGING ARRAY SYSTEM .........................................129
7. 1. Design and simulation .................................................................................................... 129
7.1.1 Microbolometer based on metamaterial structure...................................................... 129
7.1.2 Beam focusing element- dielectric lens ..................................................................... 136
7. 2. Imaging array layout design ........................................................................................... 139
7. 3. Fabrication ...................................................................................................................... 141
7. 4. Experimental measurement setups.................................................................................. 144
7. 5. Measurement results ....................................................................................................... 147
7. 6. Further improving NEP of the bolometer ....................................................................... 154
7.6.1 NEP analysis .............................................................................................................. 154
7.6.2 Miniaturized metamaterial (U-shape) bolometer ....................................................... 157
7.6.3 Miniaturized metamaterial (hybrid serpentine/spiral) bolometer .............................. 162
7.6.4 Fabrication ................................................................................................................. 164
7.6.5 Measurement .............................................................................................................. 166

CHAPTER 8. CONCLUSIONS AND FUTURE WORK ................................168
viii

8. 1. Conclusions ..................................................................................................................... 168
8. 2. Future work ..................................................................................................................... 171
8.2.1 Genetic Algorithm ..................................................................................................... 171
8.2.2 Thermal loading ......................................................................................................... 172
8.2.3 Readout circuits ......................................................................................................... 173
8.2.4 Temperature Coefficient of Resistivity (TCR) .......................................................... 174
8.2.5 THz spectroscopy....................................................................................................... 175

APPENDICES ......................................................................................................177
APPENDIX A. Operative Microfabrication Procedures ……….……………………………178
APPENDIX B. Indicator values of a W-band BWO to set the operating frequency...………182

BIBLIOGRAPHY ................................................................................................183

ix

LIST OF TABLES
Table 1.1 The properties of heterodyne detecting components ...................................................... 7
Table 1.2 Summary of direct detecting sensing elements.............................................................. 8
Table 3.1 Properties of metals used in simulations....................................................................... 28
Table 4.1 Measured resistance of proposed bolometers ............................................................... 72
Table 5.1 Physical dimensions of the bulls-eye plasmonic-assited antenna ................................. 88
Table 6.1 Physical dimensions of extended hemispherical lenses .............................................. 109
Table 7.1 Dimensions of a unitcell of designed microbolometers.............................................. 131
Table 7.2. NEP calculated parameters ........................................................................................ 148
Table 7.3 Physical dimensions of the miniaturized (U-shape) bolometer .................................. 157
Table 8.1 Temperature coefficient of resistance (TCR) of materials ......................................... 175
Table A.1 Set parameters of a RIE .….…………………………………………………………181
Table B.1 Indicator values of a BWO..…………………....……………………………………182

x

LIST OF FIGURES
Figure 1.1 The simplified schematics of THz imaging systems; top: passive imaging system,
bottom: active imaging system (reflection mode). ......................................................................... 5
Figure 1.2 The operating principle of the bolometer [26]. ............................................................. 9
Figure 1.3 The layout and block diagram of bolometer biasing circuit. ....................................... 10
Figure 1.4 Photograph and system layout of 120 element antenna-coupled microbolometer focal
plane array [31]. ............................................................................................................................ 13
Figure 1.5 Proposed bolometer detector utilizing a direct metamaterial-inspired absorber
replacing a conventional antenna element based pixel. ................................................................ 14
Figure 2.1 FSS radiated by plane wave propagating in k0 direction. ............................................ 21
Figure 3.1 The geometries of proposed metamaterial structures (a): Cross, Circle, and
Minkowski (from left), and analysis method of Floquet theorem (b)........................................... 26
Figure 3.2 The simulation results of skin effect and permeability, (a) simulation schematic, (b)
different metals, (c) the reflection of Ni changing the thickness, and (d) the reflection of various
permeability. ................................................................................................................................. 30
Figure 3.3 Waveguide transmission simulation with different metals. ....................................... 32
Figure 3.4 Simulation results(using Floquet’s port) of 0.2/0.4 THz cross dipole absorbers with
various incident angles (θ); (a),(b) Return loss/Absorption of 0.2/0.4 THz cross absorber, (c),(d)
Field distribution(E-field)/surface current of a cross absorber. .................................................... 36
Figure 3.5 Simulation results of a 0.2/0.4 THz circle absorber with various incident angles; (a),(b)
Return loss/Absorption of 0.2/0.4 THz circle absorber, (c),(d) Field distribution(E-field)/surface
current of a circle absorber. .......................................................................................................... 38
Figure 3.6 Simulation results of a THz Minkowski absorber with various incident angles (θ); (a),
(b) Return loss/Absorption; (c), (d) Field distribution (E-field)/surface current. ......................... 41
Figure 3.7 Simulated absorption results of double stacked cross absorbers, (a) double layers of
0.2 THz absorbers, (b) double layers of 0.4 THz absorbers, and (c) 0.2/0.4 THz absorbers hybrid
configuration. ................................................................................................................................ 44
Figure 3.8 Simulated absorption results of double stacked circle absorbers, (a) double layers of
0.2 THz absorbers, (b) double layers of 0.4 THz absorbers, and (c) 0.2/0.4 THz absorbers hybrid
configuration. ................................................................................................................................ 47
Figure 3.9 Fabrication process flow of THz absorbers. ............................................................... 49

xi

Figure 3.10 Photomicrographs of fabricated absorber structures: cross and circle using Ni on
LCP substrate. ............................................................................................................................... 50
Figure 3.11 Block diagram of the reflection and transmission measurement method.................. 52
Figure 3.12 Reflection measurement setup of various incident angles. ...................................... 53
Figure 3.13 Measured transmission spectra and absorption of the 0.2 THz cross type single layer
absorber, (a) time domain measurement result, (b) calculated frequency domain spectra and
absorption, and (c) comparison measured results to simulations with various incident angles. .. 55
Figure 3.14 Measured transmission spectra and absorption of the 0.2 THz circle type single layer
absorber. ........................................................................................................................................ 57
Figure 3.15 Measured frequency response (a) and absorption (b) of a 0.2 THz cross absorber
using waveguide measurement. .................................................................................................... 59
Figure 3.16 Measured response of a 0.2 THz double layer structure: (a) reference and sample
signal, (b) the absorption............................................................................................................... 61
Figure 3.17 Measured response of 0.2/0.4 THz cross absorber double layer structure: (a)
Measured frequency response, (b) the absorption. ....................................................................... 62
Figure 4.1 The layouts and EM field distribution of the bolometers with meta-inspired absorbers.
....................................................................................................................................................... 65
Figure 4.2 Simulation results of a 0.1/0.2 THz Minkowski absorber with various incident angles.
....................................................................................................................................................... 66
Figure 4.3 Fabrication process of absorbers designed by cross and Minkowski structures. ........ 69
Figure 4.4 Measured absorptance of proposed bolometers. ......................................................... 71
Figure 4.5 Measurement setup and measured result of resistivity temperature coefficient of
nickel (Ni). .................................................................................................................................... 73
Figure 4.6 Quasi-optical measuring schematics of the bolometers using a high frequency source.
....................................................................................................................................................... 74
Figure 4.7 Measured results of normalized resistance variation of the bolometer, (a)
Transmission mode, and (b) Reflection mode. ............................................................................. 75
Figure 5.1 HFSS simulation result of Electromagnetic field distribution in the bulls eye structure.
....................................................................................................................................................... 82
Figure 5.2 Simulated transverse electric field by FDTD. ............................................................. 83

xii

Figure 5.3 The simulation results of the bulls eye structure in time and frequency domain using
FDTD. (a) Time-domain transmitted signal, (b) frequency domain transmitted signal, (c)
transmitted signal (dB), and (d) incident Gaussian signal. ........................................................... 85
Figure 5.4 Schematics of the bulls-eye spoof plasmon antenna. .................................................. 87
Figure 5.5 Schematic of a Minkowski fractal that forms a unit cell of a bolometer (Shown are the
field strengths at resonance frequency)......................................................................................... 88
Figure 5.6 Photograph of the Bullseye Plasmonic-assisted antenna (100 GHz design). .............. 89
Figure 5.7 Photograph of a Minkowski fractal type metamaterial bolometer. ............................. 90
Figure 5.8 . Block diagram and photograph of the time domain measurement setup. ................. 91
Figure 5.9 Configurations of the frequency domain measurement setup. .................................... 92
Figure 5.10 Measured results of the bulls-eye plasmonic-assisted antenna using the THz time
domain measurement system. ....................................................................................................... 93
Figure 5.11 (a) Transmitted signal through the bulls-eye plasmonic-assisted antenna
characterizing at 96 GHz (-45.3 dBm); (b) transmitted signal through a reference pin-hole (-57.3
dBm). Noise floor: -71 dBm at BW: 1 GHz. ................................................................................ 94
Figure 5.12 Measured signal to noise ratio (SNR) of the bulls eye structure relative to the pin
hole................................................................................................................................................ 95
Figure 5.13 Diagram of the measurement setup for Image array. ................................................ 96
Figure 5.14 Measured resistance deviation of the bolometer detector element. ........................... 97
Figure 5.15 FDTD simulation results of a plastic (acrylic) bulls-eye antenna: (a) E-field
distribution, (b) Transmitted signals. ............................................................................................ 99
Figure 5.16 FEM (HFSS) simulation result of E-field distribution in the plastic bulls eye antenna.
..................................................................................................................................................... 100
Figure 5.17 Photographs of a fabricated plastic bulls-eye antenna. ........................................... 100
Figure 5.18 Photographs of measurement setup of a plastic bulls-eye antenna. ........................ 102
Figure 5.19 Measured received power of a metallic pinhole and plastic (ABS) pinhole. .......... 103
Figure 5.20 Measured received power of a plastic (ABS) pinhole and plastic bulls-eye antenna.
..................................................................................................................................................... 104
Figure 6.1 The schematics of proposed dielectric lenses: (a) extended hemispherical, (b) hypohemispherical. ............................................................................................................................. 106
xiii

Figure 6.2 Results of incident wave field analysis for extended hemispherical lenses (a) D = 1.6
× λ0, (b) D = 5 × λ0, and (c) hypo-hemispherical lens. ............................................................... 110
Figure 6.3 Simulated near field (E-field) radiation patterns of extended hemispherical lenses: (a)
D = 1.6 × λ0, (b) D = 5 × λ0, and (c) hypo-hemispherical lens. .................................................. 111
Figure 6.4 Simulated far field pattern of the two extended hemispherical lenses: (a) D = 5 × λ0, (b)
D = 1.6 × λ0, and (c) hypo-hemispherical lens. .......................................................................... 113
Figure 6.5 The electric far-field radiation pattern of 5 × λ0 (a) and 1.6 × λ0 (b) lenses. ............. 115
Figure 6.6 Photographs of fabricated THz lenses (left: lens array, right: microphotographs of lens
surface)........................................................................................................................................ 118
Figure 6.7 Block diagram of far-field measurement setup. ........................................................ 119
Figure 6.8 Block diagram and a photo of planar scanning for near-field measurement setup. .. 121
Figure 6.9 The (a) dielectric constant and (b) loss tangent of ABS material used in the fabrication
of lenses. ..................................................................................................................................... 123
Figure 6.10 Measured results of Near-field planar scanning: (a) Schematics of scanning, (b)
Measured intensity plot across the lens array (across two lenses). ............................................. 124
Figure 6.11 Far-field measurement results of extended hemispherical lenses; (a) D=5 × λ0, (b)
D=10 × λ0. ................................................................................................................................... 126
Figure 6.12 Measured transmission coefficients of two different lenses with the same diameter.
..................................................................................................................................................... 127
Figure 7.1 The geometry of SRR structure, (a) a modified SRR unitcell, (b) a straight connection
unitcell, and (c) a zigzag connection array of unitcells. ............................................................. 130
Figure 7.2 Simulation results (reflectivity and absorption) of a proposed unit cell of imaging
array, (a) V- and H-polarization of 0.1 THz unit cell and (b) V- and H-polarization of 0.4 THz
unitcell......................................................................................................................................... 132
Figure 7.3 Surface current distributions of each excited mode, (a) Magnitude and (b) Vector of
surface currents, V- and H-pol (from up and down). .................................................................. 133
Figure 7.4 Schematics and simulation results of designed dielectric lens, (a) layout, (b) far-field
radiation (E-field), (c) gain, and (d) radiation pattern of the lens. .............................................. 137
Figure 7.5 Layouts of 0.1 THz imaging array pixels, (a) configuration of a pixel, (b) the straight
connected imaging array, and (c) the zigzag connected imaging array. ..................................... 140
Figure 7.6 Microfabrication process of a proposed THz imaging array. .................................... 142
Figure 7.7 Microphotographs of manufactured THz imaging array. .......................................... 143
xiv

Figure 7.8 Measurement setup block diagrams, (a) Absorption measurement setup, and (b)
Imaging measurement setup. ...................................................................................................... 145
Figure 7.9 Measuring setup of noise equivalent power. ............................................................. 146
Figure 7.10 Measured result of power absorption of THz imaging array. ................................. 147
Figure 7.11 Measured resistance versus (a) current and (b) power. ........................................... 149
Figure 7.12 Measurement setup (a) and scanned result (b) of punched holes on the metallic sheet.
..................................................................................................................................................... 151
Figure 7.13 Scanning layout and results of concealed S-shaped metal, (a) scanning layout, (b)
scanned metallic S-shape pendant, (c) and (d) scanned results with different scanning
resolution.(set raster scanned resolution: (c) 4mm/step and (d) 2mm/step). .............................. 152
Figure 7.14 Analysis result of NEP, (a) dimension dependent and (b) temperature dependent. 156
Figure 7.15 Schematics of the miniaturizing (U-shape) bolometer structure, (a) single arm, (b)
double arms, (c) triple arms, and (d) quad arms. ........................................................................ 158
Figure 7.16 Simulated results of a miniaturized (U-shape) absorber (V-pol.), (a) reflectivity and
(b) absorption. ............................................................................................................................. 159
Figure 7.17 Simulated results of a modified absorber (H-pol.), (a) reflectivity and (b) absorption
ratio. ............................................................................................................................................ 160
Figure 7.18 Simulated results of a proposed serpentine absorber, (a) schematic, (b) reflectivity,
and (c) absorption ratio. .............................................................................................................. 162
Figure 7.19 Mask layout (a) and photographs of fabricated bolometers (b). ............................. 164
Figure 7.20 Layout and photographs of a lens array (a) and E-field simulation (b)................... 165
Figure 7.21 Measurement setup for the assembly of a bolometer and a lens array. ................... 166
Figure 7.22 Measured results of resistance variation.................................................................. 167
Figure 8.1 Various schematic of unicells generated from GA [94]. ........................................... 172
Figure 8.2 Conceptual pictures of thermal isolation, (a) cavity, and (b) integration a bolometer
with a readout circuit. ................................................................................................................. 173
Figure 8.3 Proposed system level assembling methods and readout circuit. .............................. 174
Figure 8.4 Simulation result for a muti-band unitcell, (a) layout of a unicell and (b) the
preliminary simulation result of absorption. ............................................................................... 176
Figure A.1 The etching rate of the copper etchant ......………………………………...……….180
xv

Figure A.2 Nordson March RIE-1701…………...……………………………………..………181

xvi

CHAPTER 1. THZ IMAGING ARRAY TECHNOLOGY
1. 1. Introduction
Over the last decade significant interest has grown in the area of terahertz (THz). The
THz electromagnetic spectrum lies between millimeter wave (mmWave) and infrared (IR) band.
Broadly, it is defined by the frequency range of 0.1 to 10 THz with corresponding free space
wavelength (λ0) between 3 mm to 30 µm [1]. Classically, another definition is that, the
frequency range is applied to 300 GHz to 3 THz (λ0= 1 mm to 1 µm) [2]. THz frequency range
has several advantages; in comparison to mmWaves; 1) higher resolutions, 2) miniaturization of
components sizes (antenna, passive and active circuits), in comparison to IR and X rays; 1) nonionizing radiation (unlike X-rays), 2) better transmission under poor visibility environments such
as smoke, fog, clouds, and sandstorms [3]. Based on these unique characteristics, a large number
of applications have evolved in this spectral range. In particular, THz imaging systems are being
developed for applications including concealed weapon and explosive detection for security and
the radiation spectral of interstellar for radio astronomy [2, 4, 5]. THz imaging systems are also
being investigated as a complement to existing screening methods such as X-rays, magnetic
resonance imaging (MRI) and ultrasonography for biomedical applications [6]. In particular THz
imaging systems are attractive for screening skin cancer and tooth decay where X-rays are
commonly used. Use of THz imaging will reduce exposure of patients to high energy X-ray
radiation.
For communications, despite the high attenuation of THz radiation through atmosphere
in compare to RF frequencies, there is growing interest its use for high bandwidth

1

communication for short range point to point communications [8]. This includes communications
within a building or from a computer to computer. Using THz carrier frequency, the usable
bandwidth can be increased manifold, it can help transmit enormous amount of data (5 % of 300
GHz=15 GHz * 1 bps/Hz=15 Gbps at QPSK) and for military applications it enhances the
jamming capability using conventional spread spectrum techniques. Furthermore, due to natural
small wavelength, the physical size of sub-system components can be significantly reduced in
size and thus saving cost and reducing weight of most systems. The components needed for THz
systems are non-existent, and the ones that are commercially available are cost-prohibitive and
limited in their functions. Similar to RF and millimeter wave systems, the components needed
for most system (e.g. antenna, phase shifters, polarizers, and so on) are similar in nature. Thus,
THz component will find applications in a broad range of systems. In order to narrow down the
focus, this research focuses on components that are needed in the design of a THz imaging
system. In particular an imaging system built using bolometer detectors. Components
demonstrated for imaging array will find utilization in a host of other THz circuit designs (e.g.,
communication links).
As mentioned above, THz imaging is being studied for many applications ranging from
security to medical imaging. Thus, there is significant interest in the development of THz
imaging systems [9-13]. There are various approaches to designing a THz system, and one
approach is through the use of a focal plane array (FPA) chip. This approach allows in the use of
existing micro- and nano-fabrication infrastructure utilized in the manufacture of high
performance front-end RF components.
The FPA chip is an integral part of a THz imaging system. In particular, THz
components designed from periodic structures are studied. Components include THz detectors,

2

beam focusing elements and antenna. Detector elements that work similar to a bolometer
detector element that is commonly used in the infrared region are studied. Beam focusing
elements includes lens array designs and also beam focusing through surface plasmonics and
dielectric lenses. All of these components are designed to operate in the THz frequency region (>
100 GHz). Also, all of proposed components are designed, fabricated, and tested under this study.

1. 2. THz Imaging system
In this section, existing THz imaging systems that have been presented in literature
and their respective sub-components are presented. A THz imaging system can fundamentally be
divided into two separate groups: i) passive and ii) active imaging system. Passive imaging
systems directly detect the natural radiation from the objects or the reflection from the
environment; the concept is analogous to radiometry to measure the electromagnetic radiation to
be characterized by difference in temperature and emissivity [14-16]. The architecture of passive
imaging systems (depicted in Figure1.1) is quite similar to a visible camera structure that
consists of detectors, focusing elements. In passive imaging system, there are two methods to
monitor the THz radiation: heterodyne detecting and direct detecting. For heterodyne detection,
the semiconductor (Schottky diode mixer), superconductor (cryogenic cooling), and hot electron
bolometers (HEB) have been extensively investigated over the past decade [2, 17-19]. One of the
basic configurations of direct detection approaches comprises of bolometers and an antenna
element. The antenna elements directly or indirectly couple to the bolometer element providing
an impedance match between the bolometer element and the free space. Although coupling
efficiency is enhanced but the antenna element thermally loads the detector and thus reducing
sensitivity. The bolometer detectors are also broadband in nature and thus utilized where

3

frequency resolution is not required. Frequency bandwidth is limited by the antenna element
utilized.
An active imaging system has a source to illuminate the object with the
electromagnetic wave and subsequently detect the reflected electromagnetic wave. These are, in
general, narrow band systems because narrow band generation of radiation is simpler.
Furthermore, sensitivity can be enhanced through heterodyning. This operating principle is
similar to that of radars. This imaging system has been in the spotlight for indoor security
application where the detection range is limited [20] and also for environment monitoring for
hazardous plumes during a fire. Practically, in order to achieve sufficient signal to noise ratio
(SNR) to detect an object in an indoor environment, where the background noise level is close to
natural radiation level from the sample, the active imaging system shows better performance than
the passive imaging system. To be specific, compared to the passive system, although both of the
fundamental structure of power detecting are same, the active imaging system can increase the
capabilities such as higher sensitivity. Due to this characteristic, the active imaging system has
been used in remote scanning applications.
In a passive imaging system, the system has to be designed to be wide band to capture
ambient signal over a wide frequency range to achieve good detection capability (i.e., capture
black body radiation reflected from the objects over a wide spectral range). On the other hand, an
active imaging system is typically made to be narrow band as the illumination signal used is
generally at a fixed frequency or over a narrow band. Thus, the components to be designed
should cater specifically to this need.

4

Figure 1.1 The simplified schematics of THz imaging systems; top: passive imaging system,
bottom: active imaging system (reflection mode).

5

1. 3. Components for THz imaging system
In this section, two essential sub-components for THz imaging systems are introduced:
detectors and focal plane array.
1.3.1THz detectors
THz detectors play a significant role in order to measure the incident signal in a THz
imaging system. THz detectors can be divided into two categories: heterodyne and direct
detectors [2]. Heterodyne detectors are the most common THz detectors using semiconductors
and superconductors. In a heterodyne detection approach, both a RF source and a local oscillator
(LO) are needed. Detection is carried out by mixing the RF and LO signals. The key advantage
of this approach is that phase information can be preserved. Direct detectors have been studied in
order to replace heterodyne detection methods in imaging arrays where phase information is not
necessary. Also, this is because a system based on heterodyne detectors consumes significantly
more total DC power than a system based on a direct detector. Furthermore, the design of a
heterodyne imaging system where a large number of pixels are utilized becomes difficult to
design as the local oscillator (LO) power distribution on a chip becomes challenging.
Illumination of LO can be utilized; however, for a large number of pixels the total LO power
required will be significantly high. Typically, LO power of 3 – 10 dBm per pixel is required for
good sensitivity.
For heterodyne detection, the semiconductor (Schottky diode mixer), superconductor
(cryogenic cooling), and hot electron bolometers (HEB) have been widely used [2, 17-19].
Conventionally, the Schottky diode mixer has been utilized in applications where the sensitivity
of room-temperature is acceptable. Also, it offers sufficient radiometric sensitivities near 0.5 K at
0.5 THz and 0.5 K at 2.5 THz. In superconducting detectors, in order to achieve high sensitivity
6

cryo-cooling is required which makes the overall system large and bulky. These detectors have
been developed based on the Josephson-effect, superconductor-semiconductor barriers,
superconductor-insulator-superconductor (SIS) tunnel junction mixer, and bolometers [2, 19].
Hot electron bolometer (HEB) mixer is an alternative to THz SIS mixers that can work at very
high speeds owing to fast phonon or electron diffusion cooling [21]. Recent HEB mixers
essentially based on tiny microbridges of niobium (Nb), NbN, and aluminum (Al) that respond
thermally to THz radiation [22]. Heterodyne approach provides significant advantage of the
ability to track the IF up to several gigahertz with the fast voltage responsivity (picoseconds
range) [23]. Table 1 summarizes the properties of the heterodyne semiconductor,
superconductor, and hot electron bolometers.

Table 1.1 The properties of heterodyne detecting components
Semiconductors
(Schottky diode mixer)

Advantages

Superconductors
(SIS mixer)

Hot electron bolometers
(HEB mixer)

 High SNR
 Low LO power (μW)
 Fast
voltage
responsivity (pS range)
 Working at room temp.  High
sensitivity(0.05K/GHz)

Low LO power (nW
 High sensitivity
 High
operating range)
frequency
 Schematics
and
materials for THz
 High LO power(mW)

Disadvantages

 Earth science
Applications  Planetary observation
 Plasma diagnostics

 Superconductor:
 Noise
performance
NbN/AlN/NbN
related to the material
 Cryogenic
cooling critical temperature and
operating temperature
system needed
 Earth science
 Space observation

7

 Space observation

In the case of direct detectors, there is a rising need to replace the heterodyne detecting
in some applications that necessitate relatively low spectral resolution compared to the ultrahigh
resolution of heterodyne detectors and also to reduce overall DC power needed for detection
[2,6]. There have been several types of detection approaches used including GaAs-Schottky
diodes with antenna coupled, bolometers using thermal absorbing materials such as bismuth (Bi)
and vanadium oxide (VOx), compound bolometers consist of thermometer or readout circuits
integrated with the radiation absorber, microbolometers with focusing elements to couple
incident power, acoustic bolometer, a fast calorimeter, etc. [2,6]. However, these detectors based
on the bolometer have challenging problems to solve including a calibration (mode matching)
problem in antenna coupled detectors (diode and bolometers) and slow response time (the order
of seconds) of an acoustic bolometer and a fast calorimeter [6]. In order to resolve this problem,
the cooled detector based on He-cooled Si, Ge, or InSb bolometers have been used for THz
detection.
Table 1.2 Summary of direct detecting sensing elements
Schotty diode antenna
coupled
Advantages

 Room temperature
 Fast response

Bolometer

Cooled detector

 Direct thermal
 Fast response time
absorption
(microsecond scale)
 Simple readout circuitry

 Circuit biasing (for non-  Thermal loading from  Cryogenic cooling
antenna
Disadvantages zero bias diodes)
 Cryogenic cooled
readout circuits
 Expensive substrate
 Calibration
 Slow response time
 Focal plane array
Applications  Security (passive and active imaging system)
 Non-destructive evaluation (active imaging system)
 Biomedical imaging and microscopy

8

Here, more detail information of detectors is presented, starting with a bolometer.
Bolometers were introduced by S.P. Langley in 1878. His invention, the bolometer, works as a
radiant-heat

detector

that

is

very

sensitive

to

temperature

variation

(temperature

resolution=0.00001 °C) [24]. The operating concept of the bolometer is to measure the electrical
resistance of the material (Bi, VOx) responds to the incident power of electromagnetic waves.
Fig 1-2 demonstrates the general scheme of the bolometer [25]. Absorbers are normally realized
using thin metal/semiconductor film. The thermistor is in direct contacts with the thin film. In
this configuration, the thermistor is in direct thermal contact and thus decreases thermal
capacitance [26]. Second, there is a detector using photoconductive response with certain
composite material, Gallium doped Germanium (Ge:Ga). It has a distinct characteristic that the
cut-off wavelength is roughly 120 µm under the unstressed configuration and 4.2 k; however,
when the stress is induced to a detector crystal layers, the wavelength of it increases up to 200
µm [27]. The stress can be induced through thermal loading. This type of detector has been
utilized in astronomy and laboratory applications.

Figure 1.2 The operating principle of the bolometer [26].

9

In order to analyze the bolometer operating mechanism, the basic theory bolometer is
introduced. A bolometer is conventionally composed of an absorber and a thermometer of heat
capacity C, connected by a thermal conductance G (substrate and circumstance of a bolometer),
to a heat sink retained at a certain fixed temperature T0 in shown as Figure 1.3 (a) [28]. The
power of incident wave, P is transferred to heat in the absorber. This causes the temperature to

, until the radiation power going to the absorber is identical to the

rise,

power running into the heat sink through the weak thermal linkage. Figure 1.3 (b) shows a
typical bolometer measurement setup. The bolometer element is placed in series with a large
stable resistor (low noise and low TCR) and biased using a DC voltage source [28, 29].

(a)

(b)

Figure 1.3 The layout and block diagram of bolometer biasing circuit.
From the bias source Vbias and the load resistor RL, the current I flows constantly runs
through the bolometer. The incident power causes the bolometer temperature (T) to increase and
increasing its resistance value. This relation can express the equation,
10

1-(1)
T, the temperature rise has an influence on change in the resistance of the bolometer and
thus the voltage, VB across a bolometer changes which is measured using a readout circuitry [28,
29].
Conventionally, there are several critical parameters that affect the performance of a
bolometer. The performance is described through key parameters such as noise equivalent power
(NEP), noise equivalent temperature difference (NEΔT), time constant (τ), and detectivity (D*).
NEP is a fundamental parameter, it describes the sensitivity of a bolometer and is the minimum
detectable power (signal to noise ratio SNR of 1) over a bandwidth of 1 Hz [29,30].
Theoretically, the NEP per unit Hz bandwidth of a bolometer is defined as [30]

√

1-(2)

2

-8

Where, A = area of a bolometer (m ), σ = Stefan-Boltzmann constant (=5.67×10
2

4

J/(m ∙S∙K )), k = Boltzmann constant, T = temperature, ε = emissivity. These parameters can be
optimized to achieve high sensitivity, a low NEP value is desired (i.e., high sensitivity). To
achieve high sensitivity, small bolometer area, low-thermal conductivity, high absorptivity and
strong temperature dependence of the bolometer film is desired. From thermal point of view,
sensitivity can be express as NEΔT,

√

11

1-(3)

Where, B is the system bandwidth and dP/dT is the change in power on the bolometer per
unit change in temperature in the spectral band. The detectivity and responsivity can be
expressed [28, 30]

√

1-(4)

√
The response time (τ) of a bolometer is given by

1-(5)

From this equation, we can recognize the relation between the heat capacity and thermal
conductance. In order to increase response time, thermal capacity C should be made small and
thermal conductance to the substrate G should be made large. However, increase in thermal
conductance leads to poor NEP. Thus, there is an optimum solution between sensitivity and
response time.
1.3.2 THz focal plane arrays
THz focal plane array is the use of the microbolometer (or other detection elements)
arrays which are fabricated by integrating the microbolometers with planar antennas, for
instance, bow-tie antenna and log periodic antenna onto a common substrate. As an example, in
[31], a 120 element antenna coupled microbolometer FPA for concealed weapons detection was
introduced. Here niobium (Nb) films are utilized for the bolometer element and slot ring antenna
for coupling the incoming radiation to these elements. For applications in an indoor environment,
the operating signal level is low because of natural radiation level is close to the background
noise power level. In this case, the active imaging system is preferred for higher sensitivity over
the passive imaging systems. Active imaging is achieved by including an illuminator along with
12

these detection elements. The FPA introduced in [31] is manufactured on a high resistivity
silicon wafer (D=75mm), and integrates 120 Nb microbolometers coupled to slot-ring antennas,
see Figure 1.4. The operating mechanism is that the antennas focus the incident sub-THz
radiation and a voltage difference is produced across the Nb bolometer depending on the amount
of incident power. In this system, in order to generate the signal, three IMPATT pulsed noise
oscillators are used (rated power = 1W, frequency=95GHz) configured to illuminate the scene
from three directions. Illumination from multiple angles assists signal detection regardless of the
orientation of typical targets.
As discussed here, significant progress has been made in the design and fabrication of
bolometer based THz imaging system. However, there are many advances that need to be made
to make this technology viable to everyday use. This includes: i) enhancing sensitivity of the
bolometer element, ii) integration of micro lens for further enhancement of sensitivity. Also,
there is a need for CMOS based readout circuitry that needs to accompany the individual pixels
and thus the processes developed should be compatible with CMOS fabrication or CMOS post
processing.

Figure 1.4 Photograph and system layout of 120 element antenna-coupled microbolometer focal
plane array [31].
13

1. 4. Goals of the dissertation
First, detectors based on frequency dependent periodic structures are investigated.
Compared to other bolometer design, in place of an antenna element, periodic structures are
incorporated to enhance sensitivity, see Figure 1.5. Furthermore, skin-effect losses are taken
advantage of to further enhance sensitivity. Thus, use of a lossy metal coupled with patterned
structures is implemented in THz direct detecting sensors. The absorbing film is made from a
periodic metallization, which combines the loss of the metal due to surface resistance with the
field enhancement near the structural resonance to achieve absorption losses similar to a
bolometer. Existing microbolometers have been replaced with a simple fabrication and low cost
metamaterial-inspired bolometer.

Micro-optics

t

Absorber
Back Reflector
Absorber

Conventional Approach

Proposed Approach

Figure 1.5 Proposed bolometer detector utilizing a direct metamaterial-inspired absorber
replacing a conventional antenna element based pixel.

14

Second, microbolometers have been used in conjunction with various antenna designs
to achieve improved coupling efficiency. The electromagnetic energy heats the microbolometer,
altering its electrical resistance, thus altering the applied current or voltage. These systems boast
small length scales and heightened sensitivity to small changes in the incident electromagnetic
energy. However, the coupling of these two systems can lead to unwanted thermal loading,
negatively affecting the overall sensitivity of the sensor. Also, small size bolometers are
desirable to improve signal to noise ratio. Thus, a sensor which not only captures a large amount
of incident electromagnetic energy, but also efficiently converts this to a usable electrical signal,
is desirable. This takes advantage of not only the thermal sensitivity of the bolometer, but also
improves its coupling efficiency with the frequency selective surface. The focal plane array
(FPA) forms the heart of the THz imaging system. The challenge is in the fabrication of beam
focusing elements at the individual pixel element level. For current imaging systems, built using
single pixel elements, silicon micro lenses are used. However, Si lenses are made from high
resistivity Si that is expensive and difficult to fabricate in a large array format on a single wafer.
A simple fabrication that allows simultaneous fabrication of multiple lenses in an array format is
desirable which will help reduce complexity in assembly. In this research, in order to realize the
focal plane array (FPA) with improving coupling efficiency and integrating with the proposed
bolometers, surface plasmon antenna and micro-lens array are studied.
Finally, in view of system level, the way to integrate all of proposed components such
as the detector using periodic structures and beam focusing antenna together and characterize the
system level performance has been carried out. In order to measure the system performance,
diverse measuring methods such as quasi-optic measurement, time-domain measurements has

15

been considered. Through this process, the performance of proposed THz imaging system has
been characterized.

16

1. 5. Overview of the dissertation
This dissertation is organized as follows:
Chapter 1 introduces THz imaging array and introduces existing approaches. Chapter 2
explains theoretical methods to analyze the proposed periodic absorbing structures using periodic
analysis method. The fundamental properties of periodic structures such as frequency selective
structure and metamaterial are studied.
Chapter 3 demonstrates several types of THz absorber based on periodic structures that
operate at THz frequency spectrum range. The detail information of design, EM simulation and
fabrication of THz absorbers are not only introduced, but also the measurement results using
time and frequency domain testing setups are explained in this chapter.
Chapter 4 describes a novel THz power detector using designed THz absorber. The
proposed power detector is demonstrated for W-band application and used in the construction of
designed periodic structures. The material properties of nickel are characterized using measuring
setup of temperature coefficient of resistance (TCR). Quasi-optic measurement has been utilized
to examine the power meter at designed operating frequency.
In Chapter 5, in order to improve the incident power intensity to the imaging sensor, the
beam focusing elements (bulls eye antenna) based on surface plasmonic are investigated. Two
well-known simulation methods, finite element method (FEM) and finite difference time-domain
(FDTD) have been utilized in the design of bulls eye structures. For the fabrication, two methods
are investigated: i) metal etching and ii) 3D plastic printing. Deep trench wet-etching of copper
(Cu) is introduced and utilized instead of the conventional micromachining techniques. A new
plastic based bulls-eye structure using three-dimensional printing is investigated. For antenna

17

measurement, the relative gain of the bull’s eye structure is measured relative to pinhole in a Cu
substrate.
Chapter 6 presents different types of beam focusing elements (dielectric lens) that can
be used to improve the sensitivity of the detector elements. A novel approach to integrate large
number of lens element at the wafer-level is introduced and demonstrated. Detailed simulations
using HFSS is carried out for a range of dielectric lens designs. Simulations are carried out for
near-field and far-field radiation pattern analysis. Using similar setup as utilized for the
characterization of bull’s eye structure, the micro-lenses are characterized.
Chapter 7 introduces a novel THz imaging array that integrates the above elements
together to fabricate focal plane arrays (FPAs). This includes the integration of metamaterial
absorber structure along with micro-optics at the wafer-level. To miniaturize the size of the pixel
elements to enhance sensitivity, a range of metamaterial inspired structures are designed,
fabricated and characterized. An imaging array is demonstrated by imaging a concealed object in
a paper envelope.
Chapter 8 concludes the finding of this research work, and future research topics and
potential applications using a THz imaging array are discussed.
.

18

CHAPTER 2. PERIODIC STRUCTURES: FREQUENCY SELECTIVE
AND METAMATERIAL
This chapter introduces the periodic structures utilized and their theoretical analysis.
Periodic structures such as frequency selective surfaces (FSS) and metamaterials, normally have
been used in filter application in electronic or RF devices. Unlike a conventional filter, which
normally studies only the frequency response (resonant frequency and reflection and
transmission response), the periodic structure must poses bandpass or bandstop characteristics
for varying incident angles and polarization. There is significant interest in periodic structures
(FSS) for a variety of applications such as the radome structures, reflector antenna for dual bands,
and stealth technology in order to decrease the radar cross section (RCS) of a flying object and
sea ferrying vessels [32-34]. Typically, the periodic structure is defined as an infinite assembly
of unitcells, showing the uniform structures, which may consist of one-, two-, and three
dimensional array. These schematics are typically composed of two layers; one is a metallic
patterned layer and the other is a dielectric layer forming a substrate or a superstrate. The
performance of the periodic structure is totally characterized by the schematic of the unit cell and
its arrangement in a semi-infinite pattern.

2. 1 Periodic structure
2.1.1 Properties of periodic structure
Originally, the periodic structure consists of an arrangement of either wire or slot shapes.
Wire type, the periodic structure is composed of randomly shaped conducting components,
which help guide the electric currents in a complex pattern. Compliment to these structures, the
slot shape of periodic structure consists of arbitrary shaped holes in a metallic plate, which can
19

support magnetic currents [32]. The former acts as a bandstop filter because the components are
reflective at a resonant frequency and they pass through electromagnetic waves below and above
the resonant frequency [33]. Circuit theory can be utilized to carry out simple analysis. An
equivalent circuit for a wire periodic structure is composed of a series LC pair resonant, in a
short circuit, it is that reflects the energy at the resonant frequency, alternatively, on the rest of
frequencies, the LC pair work to pass the electromagnetic wave same as open circuits. The latter
works as a bandpass filter, because the slot components are transparent at the resonant frequency
and they reflect the incident waves the rest of frequencies located on above and below the
resonant frequency. It can be analyzed same as the bandpass filter which is composed of the
inductor and capacitor that combined parallel. At the resonant frequency, the configuration of the
inductor and capacitor works an open circuit response, while the other frequencies work a short
circuit [32].
Also, many case of periodic structure use hybrid configurations that are composed of
metallic layers and dielectric layers. These schematics are typically composed of two layers; one
is a metallic patterned layer and the other is a dielectric layer forming a substrate or a superstrate.
The performance of the periodic structure is totally characterized by the schematic of the unit
cell and its arrangement in a semi-infinite pattern. A periodic array of dipole elements is depicted
in Figure 1.2. Independent of FSS shapes, the resonance mechanism of most FSS structure is
similar. In relation to resonance, consider an array of FSS structures on a dielectric substrate.
Assume a plane wave in free space impinges upon the FSS structure. The unit cell of the FSS
structure resonates at an effective electrical length of the unit cell that is a multiple of the
resonance half wavelength [33, 34]. Harmonizing to the phase front of the wave, these unit cells
have a certain phase delay. As a result, the radiation of the EM wave is scattered by individual

20

unit cells and the effective length of FSS leads to resonate the frequency associated with the
length. It provides bandstop or bandpass filter characteristics.
2.1.2 Analysis of periodic structure
Traditionally, in order to analyze the periodic structure, there are two fundamental
methods, ‘the plane wave expansion’ and ‘the mutual impedance’ to characterize the electrical
properties such as the scattering and impedance of designed periodic structures. Here is the
example to characterize the periodic structure to utilize both of ways that were mentioned above.
There is the finite array that is composed of two dimensional elements shown in Fig 2.1. In order
to analyze the periodic structure, Tsao’s and Mittra’s works [35, 36] are referred in this
dissertation. Based on these references, the analysis starts by considering the geometry of a freestanding periodic structure as shown in Fig 2.1.

Figure 2.1 FSS radiated by plane wave propagating in k0 direction.
First, in order to analyze the periodic structure, there is an assumption that the periodic
structure is infinite in size and the metal is much thinner than the wavelength [35]. In order to
21

analyze the periodic structure, there are the parameters; ⃗ , the current induced on the FSS
structure due to the incident field and ⃗⃗, the magnetic vector potential due to the induced
current (⃗). Also, assuming the time conventional is

, at the z = 0 plane, the components

associated with the transverse (z-axis) can be written as [33, 35, 36];

[

]

̿

[

]

2−(1)

where

̿

,̿
̿

,

k0= free space wavenumber

The transverse components of scattered electric field

⃗⃗

at z=0 can be obtained from

the transverse components, ⃗⃗ [33, 35].

[
[

2− (2)

]

]

Using the Fourier transform, and then the following equation is derived in the spectral
domain [35].

̃
[̃

]

] ̃̿

[

Where

22

̃
[̃

]

2− (3)

̃̿
̿

In this equation, the transform variables, a and b, are related to the x, y on the cartesian
coordinate system. When the FSS structure is stringently periodic, the induced current ⃗
be transformed to ⃗

can

is nonzero (using Fourier transform) for discrete values of the spectral

variables α and β [33, 35]. The discrete of spectral variables, αmn and βmn are related with the
Floquet harmonics for the periodic structure. The explicit expression for αmn and βmn are

2− (4)

where θ, φ are the angle of the incident plane waves .
Using the inverse transform and applying the boundary condition, it can be calculated
that the sum of the scattered field and the incident field is zero;

⃗⃗

⃗⃗

on the

conduction surface of the FSS structure. The following equation shows the relation for the
unknown induced currents on the FSS structure [33, 35, 36].

∑ ∑ [

̃
[̃

] ̃̿

]
(

23

∑∑[

̃

̃

̃
] [̃
̃

̃
[

]

]

2
2−(5)

Above equation 2-(5) can be solved with the Galerkin’s method. The unknown
induced current ⃗ is represented in relation to a set of basis function

⃗

∑

⃗⃗ [35];

⃗⃗

(
2-(6)

Where

Ci is still unknown coefficients to be calculated. Substituting 2-(6) into 2-(5)

and using ⃗⃗ as the testing functions, 2-(5) is changed the equation as the matrix form and can be
calculated the

Ci.

After calculating

Ci,

The unknown induced current ⃗ is obtained from the

equation 2-(6). The plane wave expansion is based on the scattering problem by the theory of
antenna array where receive the incident power and then reflect or transmit it. In order to analyze
this theory, here is the two-dimensional where is located in the XZ - plane.

24

CHAPTER 3. TERAHERTZ ABSORBERS
3. 1. Metamaterial structure
Conventionally, metamaterial structures are designed by placing various periodic
structures [37] to have two or three dimensional patterns in a metallic or a dielectric substrate. In
general, the metamaterial structure (unit cell) is much smaller than conventional frequency
selective structures (FSS, l/2 element) at the wavelength of wanted frequency. The operation
mechanism of metamaterial resembles the traditional FSS and both are based on the resonant
elements. Simply stated, the idea is that plane-wave illuminates an array of metamaterial
elements, thus exciting electric current on the surface of metamaterial structure. The amplitude
of the generated current depends on the strength of the coupling of energy between the incident
wave and the structure. Traditionally, on the FSS, the coupling reaches its highest level at the
half wavelength of operating frequency, however, on metamaterial, the physical size of the
structure can be reduced significantly by utilizing a variety of LC (inductive-capacitive) resonant
structures.
Previous

metamaterial-based

absorbers

are,

in

order

to

effectively

absorb

electromagnetic radiation in the terahertz frequency regime, utilize thin periodic structures
suspended above a metallic backing [38, 39]. These two metallic layers sandwich a thick
dielectric layer. The purpose of the system is to absorb incoming electromagnetic radiation at a
particular frequency regardless of polarization or incident angle. The periodic metamaterial unit
cell is chosen to correspond to roughly one half of a wavelength. This ensures resonance mode
build up across the structure. In order to demonstrate applications of metamaterials for
absorption, three-dimensional finite element method (FEM) simulation was performed.
25

Simulations are based on the Floquet theorem [32-35]. Figure 3.1 shows the unitcells structures
and the side view of the stacked layer.

Cross

Circle
(a)

Minkowski

(b)

Figure 3.1 The geometries of proposed metamaterial structures (a): Cross, Circle, and
Minkowski (from left), and analysis method of Floquet theorem (b).

26

3. 2. Skin Effect
The terminology of skin depth, normally, used in order to express the relative intensity
quantitatively to which the electromagnetic wave can penetrate into the metal conductor. The
skin effect causes the effective resistance of the conductor to increase at higher
frequencies where the skin depth is smaller, thus reducing the effective cross-section of the
conductor. In this research, we studied the effect of the metal used to make the patterns of the
terahertz metamaterial-inspired absorber. Equation 3-(1) shows the expression for skin depth (δ)
[40]. Which is dictated by conductivity () and permeability (r) of the material and frequency (f)
of operation.

√

√

3 − (1)

It is well known that the smaller the skin depth, the more lossy the metal is to a
propagating wave. From Equation 3-(1), it is clear that a small skin depth can be achieved
through the use of a large permeability or a low conductivity material. On the other hand, in
order to make lossy conductors, there are two ways to be considered. First, a metal can be forced
to be lossy by enforcing a thickness of the metal film that is thinner than its skin depth at the
desired frequency. Second, the surface impedance of a metal can be expressed the relationship
between the resistivity and skin-depth shown as equation 3-(2) [41]. In this equation, the real and
imaginary parts are the resistance and the inductance of a surface conductor. If the skin-depth is
made small, it leads to higher surface resistance with the same bulk resistivity.
3 − (2)
Leading on from these two ideas, two different studies were conducted. First, the effect
of using high-conductivity copper (Cu) and low conductivity titanium (Ti) were studied. In this
27

case, the thickness of metal was chosen to be smaller than the skin depth in order to use the high
surface resistance at resonance, and hence, increase the loss of the absorber. Second, a
ferromagnetic metal, nickel (Ni) was considered. This metal has a high permeability, albeit
unknown in THz region, which is related to skin depth. The small skin depth of nickel makes the
surface highly resistive. This high surface resistance interferes with the wave propagation, which
results in resonance absorption losses. The properties of metals related to the skin effect are
summarized in Table 3-1.
Table 3.1 Properties of metals used in simulations
Nickel

Titanium
7

Copper

6

7

Conductivity (S/m)

1.45×10

1.82×10

5.8×10

Permeability(μr)

100

1

1

0.1 THz

0.042 μm

1.16 μm

0.20 μm

0.2 THz

0.030 μm

0.80 μm

0.14 μm

0.4 THz

0.021 μm

0.57 μm

0.10 μm

Skin depth
(δ)

In order to verify these distinguished properties, the absorption characteristic of each
metal is simulated using a commercial electromagnetic simulation tool, Ansoft HFSS based on
finite element method (FEM). Using this simulation tool, the S-parameter can be produced to be
able to characterize and calculate the absorption (

|

|

|

|

) of the

absorber. Figure 3.2 (a) illustrates the scheme of the simulated dipole structure to analyze the
correlation between skin effect and metal properties. It is composed of a conductor layer, a
dielectric layer, and ground plane (a single port network due to ground plane). The simulated
dipole structure has the dimensions, in micrometers (μm), of: width (W) = 75, length (L) = 400,
and conductor thickness (t) = 0.2, respectively. Simulation results of return loss of different
metals are shown as Figure 3.2 (b). As mentioned above, the high permeability metal, Ni, has a
28

significant absorption characteristic under same conductor thickness compared to other metals
due to the thinner skin depth. Figure 3.2 (c) illustrates the simulated return loss of a Ni dipole to
analyze the absorption ratio on different thickness. The results show that thinner Ni layer absorbs
large amount of the incident power, the peak absorption is close to 0.99 at t=0.01μm. Figure 3.2
(b), (c), and (d), it is clear that the high permeability and thin conductor thickness enable to
achieve perfect absorption of the proposed absorbers due to high surface resistivity.

29

(a)

Different metal Dipoles (Ni/Cu/Ti) - TE with various incident angles

0

S11 (dB)

-5
-10
-15

Ni()
Ni()
Cu()
Cu()
Ti()
Ti()

-20
-25
160

180

200

220

240

Freq (GHz)

(b)
Figure 3.2 The simulation results of skin effect and permeability, (a) simulation schematic, (b)
different metals, (c) the reflection of Ni changing the thickness, and (d) the reflection of various
permeability.

30

Figure 3.2 (cont’d)
Different thickness of Ni - normal incident

S11 (dB)

0

-10

t=0.0001mm
t=0.0003mm
t=0.0005mm
t=0.0007mm
t=0.001mm

-20

160

180

200

220

240

Freq (GHz)

(c)
Different permeability

0

S11 (dB)

-5

-10
r=1
r=5

-15

r=10
r=21

-20

r=46
r=100

-25
160

180

200
Freq (GHz)

(d)
31

220

240

Another simulation has been carried out that the transmission properties of a rectangular
waveguide coated with different metals mentioned above. Figure 3.3 shows the simulated
transmission (S21) of rectangular waveguides. The physical size of waveguide is set as WR-4
(operating frequency: 170 to 260 GHz), W × H=1.0922 mm × 0.5461 mm. The coated thickness
of metals is 100 μm.

Transmission loss (dB)

0

-2

Ni
Cu
Ti

-4

-6

-8

-10

100

150

200

250

Frequency (GHz)
Figure 3.3 Waveguide transmission simulation with different metals.

32

300

All of the simulations show that nickel, a ferromagnetic material, has higher absorption
and wider bandwidth than the other metals considered. On the other hand, the higher
conductivity material, copper (Cu), showed the narrowest absorption bandwidth. Hence, nickel
(Ni) was chosen for the design and fabrication of absorbing structures. Nickel is also a preferred
metal for fabrication of packaging devices, because it is a low-cost material and can easily be
deposited using electroless- and electro-plating [42].

33

3. 3. Designs and simulations
3.3.1 Single layer proposed metamaterial arrays
For the actual device design and testing, 0.2 THz and 0.4 THz were chosen as the center
frequencies. The exact shapes of the metamaterial unit cell were chosen to emulate a regular
cross, circle, and Minkowski as shown in Figure 3.1. All of proposed structures were considered
to produce the polarization independent form it of incident wave. The widths (W) of each cross
arm were chosen as 75 μm and 37.5 μm, while the length (L) of each arm was chosen as 400 μm
and 200 μm, respectively. Additionally, the gap (G) between each unitcell was set as 150 μm and
75 μm. In the circle layout, the radius (R) of it was designed as 204 μm and 102 μm, the gap (G)
set 72 μm and 36 μm, separately. For Minkowski structure, the width (W) of the structure was
designed as 760 μm, the length (L) of a leg was 345 μm, and the entire physical size of unitcell
(U) is 830 μm. Likewise, the thickness of the metallic layer was chosen to be 0.5 μm and 0.25
μm, while the thickness of the dielectric was 50 μm and 25 μm, respectively. Rogers®
ULTRALAM liquid crystalline polymer (LCP, εr= 3.5 and tan δ= 0.05) was chosen as the
substrate based on its relatively good dielectric properties in the THz spectral region [43].
Figure 3.4 demonstrates the simulation results of cross-dipole absorbers using Floquet
port in HFSS. To be specific the simulation, the electromagnetic wave is excited from a port on
top side of z-axis, which is able to radiate TE/TM mode. Also, different incident angles (θ) are
considered from 0 to 90°. Then, the simulated structure can be only computed the reflection (S11)
because of a single port. From simulated the reflection, the absorption can be inverted. In Figure
4 (a) and (b), 0.2 and 0.4 THz cross absorber simulation results are shown. Reflection is plotted
on the left side and absorption is the right side with two exiting modes and changed incident
angles. In Figure 3.4 gives the reflection and absorption as different incident angles in the TM
34

and TE, respectively. Both cross absorbers, the simulated frequencies of maximum absorbing
show 0.198/0.197 THz (TE/TM mode) and 0.387/0.389 THz at a normal incident angle. The
calculated absorption of maximum absorbing frequencies is 88.3/88.1 % at 0.2 THz and 78/78.3 %
at 0.4 THz, respectively. In simulation results, the absorbing results of TE and TE mode, both are
well matched each other. It is because this scheme (cross) has a structural advantage which is
less dependent on the polarization of incident electromagnetic wave. Moreover, up to the
incident angle, θ=60° (not shown), the absorption of a cross absorber works well both of
transmitting modes (peak absorption of TE=62.3 %, TM=86 %). The simulated full width at half
maximum (FWHM) of each frequency is calculated 15.6% (at 0.2 THz) and 19.3 % (at 0.4 THz)
at normal incident angle. The E-field and surface current distributions are illustrated in Figure
3.4 (c) and (d), respectively.
Simulated results of circle absorbers are depicted in Figure 3.5. Generally, the
schematic of circle is independent from the polarization of incident wave. At normal incident
angle, the peak absorption and FWHM of TE are identical with it of TM (peak absorption=93.8 %
at 0.201 THz, 99 % at 0.395 THz).

35

Absorption of 0.2 THz Cross Absorber TE & TM with various incident angles

1.0

0
0.8
TE ()
TE ()
TM ()
TM ()

S11 (dB)

-4

0.6

0.4
-6

Absoprtion

-2

0.2
-8
0.0

-10
160

180

200

220

240

Freq (GHz)

(a)
Absorption of 0.4 THz Cross Absorber TE & TM with various incident angles

0.8

TE()
TE()
TM()
TM()

S11 (dB)

-2

0.6

0.4

-4

Absoprtion

0

0.2
-6
0.0
-8
250

300

350

400

450

500

550

Freq (GHz)

(b)
Figure 3.4 Simulation results(using Floquet’s port) of 0.2/0.4 THz cross dipole absorbers with
various incident angles (θ); (a),(b) Return loss/Absorption of 0.2/0.4 THz cross absorber, (c),(d)
Field distribution(E-field)/surface current of a cross absorber.
36

Figure 3.4 (cont’d)
Absorption of 0.4 THz Cross Absorber TE & TM with various incident angles

0.8

TE()
TE()
TM()
TM()

S11 (dB)

-2

0.6

0.4

-4
0.2
-6
0.0
-8
250

300

350

400

450

500

550

Freq (GHz)

(b)

(c) E-field

(d) Surface current

37

Absoprtion

0

In the incident angle, θ=30° of 0.4 THz absorber, the reflection of TE shows very
deeply comparing to other TE/TM simulation. However, as converts to the absorption, the value
of it was calculated as 100% absorption. The deviation between normal angle and this angle is
only 1%. Compared to the cross type absorber, the FWHM of circle absorber is shown wider
absorption bandwidth, 19.7 % at 0.2 THz and 22.6 % at 0.4 THz, respectively. Figure 3.5 (c) and
(d) demonstrate the field distribution of E-field and surface current density on the surface of the
circle absorber.

Absorption of 0.2 THz Circle Absorber TE & TM with various incident angles

1.0

0

0.8

S11 (dB)

-10

0.6

0.4

-15

Absorption

TE()
TE()
TM()
TM()

-5

0.2

0.0

-20
160

180

200

220

240

Freq (GHz)

(a)
Figure 3.5 Simulation results of a 0.2/0.4 THz circle absorber with various incident angles; (a),(b)
Return loss/Absorption of 0.2/0.4 THz circle absorber, (c),(d) Field distribution(E-field)/surface
current of a circle absorber.

38

Figure 3.5 (cont’d)
Absorption of 0.4 THz Circle Absorber TE & TM with various incident angles

1.0
0

S11 (dB)

-10
-20

0.8

0.6

-30

0.4

-40

0.2

-50

0.0

250

300

350

400

450

500

Freq (GHz)

(b)

(c)

(d)

39

550

Absorption

TE()
TE()
TM()
TM()

Figure 3.6 demonstrates simulated results of Minkowski structure that is designed to
operate at 0.2 THz. The distinct characteristic of Minkowski has multiple resonant frequencies
due to the shape [64]. The frequencies of strong absorption peaks are approximately at 0.21 THz
(main), 0.243 THz (second) and 0.074 THz (third). Maximum absorption frequency, the amount
of S11, is 15.5 dB. It is clear that the absolute density of absorption is 97.2 % at the main
frequency. Absorptions at second and third frequency are 80 % and 79 %, respectively. All of the
peaks can be utilized for absorption in incident ration. From simulation results, the FWHMs of
absorption peaks are approximately 28.6 % (0.061THz between 0.193 and 0.254 THz including
main and second peak) and 18.2 % of third peak. The bottom of Figure 3.6 shows simulated Efield and surface current distributions for the Minkowski structure. It is clear that strong E-field
appears in the edge of structure and the EM wave penetrates the space between the metallic
layers leading to desired absorption. These geometrical parameters play a significant role in
tuning the desired frequency.

40

Minkowski fractal Absorber - TE & TM with various incident angles

0
-2

S11 (dB)

-4
-6
-8
-10
TE()
TE()
TM()
TM()

-12
-14
-16
50

100

150

200

250

300

350

Freq (GHz)

(a)
Absorption of Minkowski fractal Absorber TE & TM with various incident angles

1.0

TE()
TE()
TM()
TM()

0.8

Absorption

0.6

0.4

0.2

0.0
50

100

150

200

250

300

350

Freq (GHz)

(b)
Figure 3.6 Simulation results of a THz Minkowski absorber with various incident angles (θ); (a),
(b) Return loss/Absorption; (c), (d) Field distribution (E-field)/surface current.

41

Figure 3.6 (cont’d)

(c)

(d)

42

3.3.2 Double layers of proposed metamaterial arrays
In order to increase the absorption strength and bandwidth, multi-stacking of layers was
analyzed. In the first case, two of same layers were stacked on top of each other. This approach
provides dual advantage, the inherent resonant frequency is maintained and furthermore the
interaction between these layers provide enhancement in bandwidth. In the second configuration,
two layers having different resonance frequencies are stacked on top of each other. This
arrangement provides a significant enhancement in bandwidth. Figure 3.7 summarized the
simulation results of multi-layer of a cross absorber. In Figure 3.7 (a) shows the simulated results
of double 0.2 THz cross absorbers. Compared to a single layer 0.2 THz cross absorber, the
maximum absorbing frequency shifted slightly down to 0.185 THz due to mutual coupling
between top and bottom absorber layer. With overlapping the same resonant frequency, the
operating bandwidth is higher than a single layer absorber. Specifically, the peak absorption was
analyzed as 99.9 % at 0.185 THz and the FWHM is 30.8 % (absorption: 88.3 % and FWHM:
15.6 % of a 0.2 THz cross absorber).

43

Absorption of Stacked double 0.2 THz Cross Absorbers TE & TM with various incident angles

1.0
0

0.6

-20

-30

0.4

-40

0.2
160

180

200

220

Absorption

S11 (dB)

0.8

TE ()
TE ()
TM ()
TM ()

-10

240

Freq (GHz)

(a)
0

Absorption of Stacked double 0.4 THz Cross Absorbers TE & TM with various incident angles
1.0

-5

0.8

S11 (dB)

0.6

-15
0.4

Absorption

-10

-20
TE()
TE()
TM()
TM()

-25
-30
250

0.2

0.0

300

350

400

450

Freq (GHz)

(b)
Figure 3.7 Simulated absorption results of double stacked cross absorbers, (a) double layers of 0.2
THz absorbers, (b) double layers of 0.4 THz absorbers, and (c) 0.2/0.4 THz absorbers hybrid
configuration.

44

Figure 3.7 (cont’d)

1.0

Absorption of Stacked 0.2/0.4 THz Cross Absorbers TE & TM with various incident angles

0.8

Absorption

0.6

0.4

0.2

TE()
TE()
TM()
TM()

0.0
150

200

250

300

350

400

450

Freq (GHz)

(c)

Likewise, double 0.4 THz cross absorbers show similar performances. The peak
absorption was computed as 99.7 % at 0.362 THz and the FWHM is 34%. This assembly
provides wider bandwidth and higher absorption than a single absorber. Figure 3.7 (c)
demonstrates the simulated results of a hybrid configuration that is composed of a 0.2 THz cross
and a 0.4 THz cross. In this simulation, the peak frequencies can be recognized as multiply
resonating. First peak is at 0.17 THz with 89 % of absorption and the second peak is at 0.334THz
with 78.5 % absorption. In case of The FWHM, multi peak absorptions lead to increase in the
usable FWBH significantly, for first case it is 17.6 % and 46.4 % for the second case.

45

Figure 3.8 illustrated the simulation results of multi-layer circle absorber design.
Simulated results of double 0.2 THz circle absorbers were shown as a maximum absorption
analyzed at 0.169 THz with 71.6 %, the FWHM of it was 50.4 % (usable bandwidth 0.085 THz).
On double layers of 0.4 THz, the centered frequency was simulated at the peak absorption of
80.3 % at 0.332 THz and the FWHM was 52.1 % as well. It is clear that a double stacked
absorber plays a significant role towards improving operating usable absorbing bandwidth.
However, comparing to the single 0.2 THz circle absorber, the maximum absorption ratio
reduced slightly, it is due to the spatial impedance mismatch between top and bottom layer. It
can be matched to adjust the thickness of the dielectric substrate. In Figure 3.8 (c), simulation
results of the hybrid alignment - 0.2/0.4 THz absorbers were shown. Similar to stacked hybrid
cross absorber, there are multiple absorbing resonant frequencies calculated. The peak
absorptions were simulated: first peak placed at 0.164 THz with 88.3 % absorption and second
peak shown 95.5 % at 0.361 THz. The FWHM were 25.1 % and 28.9 %, respectively.

46

0

0.8

-2

0.6

-4

0.4

-6

TE()
TE()
TM()
TM()

-8
100

150

200

Absorption

S11 (dB)

Absorption of stacked double 0.2 THz Circle Absorbers TE & TM with various incident angles

0.2

0.0
250

Freq (GHz)

(a)
Absorption of stacked double 0.4 THz Circle Absorbers TE & TM with various incident angles

0
0.8
-2

S11 (dB)

-6

0.4
TE()
TE()
TM()
TM()

-8

-10
250

300

350

400

Absorption

0.6

-4

0.2

450

Freq (GHz)

(b)
Figure 3.8 Simulated absorption results of double stacked circle absorbers, (a) double layers of
0.2 THz absorbers, (b) double layers of 0.4 THz absorbers, and (c) 0.2/0.4 THz absorbers hybrid
configuration.

47

Figure 3.8 (cont’d)

Absorption of stacked 0.2/0.4 THz Circle Absorbers TE & TM with various incident angles

1.0

Absorption

0.8

0.6

0.4
TE()
TE()
TM()
TM()

0.2

0.0
150

200

250

300
Freq (GHz)

(c)

48

350

400

450

3. 4. Fabrication
Terahertz absorbers were fabricated using a conventional micro-fabrication approach. A
liquid crystalline polymer (LCP) substrate was chosen as it shows well behaved dielectric
characteristics in the THz range. Fine metal patterns were achieved using a lift-off process. After
spin-coating and developing the photoresist (PR, Microposit® S1800® series) layer on top of
LCP, nickel was evaporated via an e-beam process to pattern the metamaterial-inspired absorbers
in the cast. Followed by removal (and lift-off) of photoresist using acetone. Figure 3.9
summarizes the process flow.

Figure 3.9 Fabrication process flow of THz absorbers.

Nickel thicknesses of 0.5, and 0.25 μm were used for 0.2 THz and 0.4 THz absorber
designs, respectively. These thicknesses are larger than the skin depth and much thinner than the

49

wavelength at these respective frequencies. Figure 3.10 shows a photomicrograph of a sample
fabricated using Ni on LCP substrate using the process outlined in Figure 3.9.

Figure 3.10 Photomicrographs of fabricated absorber structures: cross and circle using Ni on
LCP substrate.

50

3. 5. Experimental measurement
A picometrix T-ray 2000 THz time domain system (TDS) fitted with both a receiver
and transmitter was used for the measurements. This measurement system offers the sampling
period of 0.03941 ps. These results in a frequency resolution of approximately 12.5 GHz when
acquiring the Fourier transform of the received time domain signals [44]. The transmitter radiates
a collimated electromagnetic pulse beam in the direction of the receiver. Two measurement
approaches were utilized in order to test the fabricated metamaterial absorbers. These setups
were made to address many different situations emulating their use in a high frequency electronic
package. The first setup comprises an electromagnetic wave impinging normally upon the
absorber (Figure 3.11). In the past, most metamaterial absorbers have been tested in this fashion.
Some of these tests have been conducted by varying the angle of incidence. Here, only normally
incident waves are applied and tested over a wide frequency range. The success of the absorber
at particular frequencies using the first setup demonstrates the feasibility of using it to absorb
radiated waves from an electronic package. Using this setup, we experimentally measured the
metamaterial-inspired absorbers. A reference measurement is first taken with a smooth metal
plate located at the right side of the setup (1). The transmission measurement is recorded in the
presence of the plate (at position (1)) by placing the sample in between the receiver and the
transparent mirror (position (2)). The reflection measurement, on the other hand, is performed by
placing the sample in place of the metal plate facing the transmitter across from the transparency
(see Figure3.11 for more details). Also, in order to characterize the performance of different
incident angles, absorbers have been measured a variety of incident angles depicted in Figure
3.12. As the absorbers patterned metamaterial structures have been measured with the different
incident angles, absorbers have been verified the absorption depending on radiating angles.

51

(a)

(b)
Figure 3.11 Block diagram of the reflection and transmission measurement method.

52

Specifically, mounting the test sample in the path of the beam would cause part of the
beam to transmit through the test sample and part of it to reflect away from its interface. It is this
transmission and reflection information that is utilized to characterize the absorber. This
measurement demonstrates that different reflection and transmission levels will occur depending
on the characteristics of the test sample. In the case of an absorber, to minimize scattering losses,
reflection losses must be minimal. For the samples measured here, the reflected signal was very
low, thus the absorption was calculated the same equation that was used in simulations.

Figure 3.12 Reflection measurement setup of various incident angles.

53

3. 6. Measured results
3.6.1 Reflection and transmission of proposed absorbers with various incident angles
Characterized results of a 0.2 THz cross absorber have been illustrated in Figure 3.13.
Figure 3.13 (a) shows the transmission measurement and a reference signal (in the absence of
sample) on a time domain. As we know the time domain measurement, when the absorber under
test was placed in the test setup, measured results were characterized by the magnitude and delay
of the incident wave. Thus, we can analyze the electrical performance of the sample. Figure 3.13
(b) shows the calculated absorption of this sample based on these reflection and transmission
measurement results. This shows a maximum absorption of 80% and the FWHM of 14.2 %
(bandwidth of approximately 38 GHz centered at 0.269 THz). In Figure 13 (c), measured results
and numerical analysis data are demonstrated together. In simulation, the center frequency of the
designed THz cross absorber was analyzed about 0.2 THz and the FWHM was 15.6 %, however,
the measured results show 0.27 THz as the peak absorption frequency(the frequency shift was
computed as 70 GHz upper) compared to the simulation. This is because the surface roughness
of the sample which was set in the test setup up was not evenly. As holds the sample on the
metal plate, there are many air rooms located between a metal plate and a sample to obstruct the
sample to be flatly. It makes the effective length of an absorber reduced. Thus, the operating
frequency of the absorber can be resonated upper side. For the usable bandwidth, the measured
FWHM reduced 1.4 % compared to the simulation. From different incident angles, at θ=30°, the
peak absorption decreased to 68.2 % at 0.212 THz, the deviation of amount of absorption and
centered frequency was 12 % and 57GHz comparing to the normal angle. The difference of
amount of peak absorption was primarily from the larger beam-width of the source than the size
of the absorber. It caused the misalignment and extra scattering of a path between the source,
54

sample, and receiver. It can be improved to adjust the alignment and use the large metal plate to
cover the beam-width properly.
Measured results of a circle absorber are summarized in Figure 3.14. Similarly to the
cross absorber, the time domain measurement was carried out and then the frequency domain
results was computed. In frequency domain results, the circle absorber shows wide absorption
property significantly, from 0.2 THz to 0.4 THz.

Time Domain Signals
0.2
Reference(no sample)
Absorber

Received Signal Intensity(a.u)

0.15
0.1
0.05
0
-0.05
-0.1
-0.15
0

10

20

30

40

50

60

70

80

Delay(ps)

(a)

Figure 3.13 Measured transmission spectra and absorption of the 0.2 THz cross type single
layer absorber, (a) time domain measurement result, (b) calculated frequency domain spectra
and absorption, and (c) comparison measured results to simulations with various incident
angles.

55

Figure 3.13 (cont’d)

15
Reference channel
0.2 THz cross absorber

0.6

0.4

5

0.2
0
0.0
0.1

0.2

0.3

0.4

Freq (THz)

(b)
1.0

simulation ()
simulation ()
measurement ()
measurement ()

Absorption

0.8

0.6

0.4

0.2

0.0
0.1

0.2

0.3
Freq (THz)

(c)

56

0.4

Absorption

Magnitude (a.u)

10

0.8

30

1.0
Reference channel
0.2 THz circle absorber

25

0.8

0.6
15
10

0.4

5

0.2

Absorption

Magnitude (a.u)

20

0
0.2

0.3

0.4

0.0
0.5

Freq (THz)

(a)

0.8

Absorption

0.6

0.4

simu(theta=0)
simu(theta=30)
theta=0
theta=30

0.2

0.0
0.1

0.2

0.3

0.4

Freq (THz)

(b)
Figure 3.14 Measured transmission spectra and absorption of the 0.2 THz circle type single layer
absorber.

57

From calculations, the occupied bandwidth above 40 % absorption was characterized to
be from 0.25 THz to 0.393 THz. The peak absorption is 82.9 % at 0.350 THz as shown in Figure
3.14 (a). Also, the harmonics of the resonant frequency can be noted in measured results. Figure
3.14 (b) shows the comparison between simulations and measured results with various incident
angles.
3.6.2 Waveguide measurement of proposed absorbers
Frequency domain transmission data for a 0.2 THz cross absorber using the waveguide
measurement setup is provided in Figure 3.15. For this measurement, this absorber was
assembled with a back metal layer. Also shown in Figure 3.15 (a) is the reference measurement
that was characterized on a dielectric waveguide. This was carried out in the absence of the
absorber. Figure 3.15 (b) shows the calculated absorption of this sample based on these reflection
and transmission measurements. This shows the peak absorption 74.5 % at 0.212 THz and strong
overall FWHM was measured 12.2 % between 0.180 and 0.26 THz. It should be noted that the
measured center frequency is different for this sample design between the two measurement
approaches. This is due to dielectric loading making lowering frequency in the second test setup.
In this setup, the THz dielectric waveguide is touching the absorbing layer. From these
measurements, it can be stated that due to the almost zero incident angle in the waveguide
measurement setup (as compared to the normal incident angle using the transmission/reflection
measurement setup), the scattering and reflection are decreased. Hence, this resulted in an
improved absorption performance. Additionally, the maximum absorption occurs at roughly the
same frequency values in both measurement cases. From these measurements, it can also be
stated that the absorbers are expected to have strong absorption over a wide range of incident
angles.
58

12

Measured spectra of a 0.2 THz Cross absorber
on Frequency domain
waveguide
single 0.2THz cross absorber

Magnitude (a.u.)

10
8
6
4
2
0
0.2

0.3

0.4

Frequency (THz)

(a)
0.8

Absorption of a 0.2 THz Cross absorber
using Waveguide measuremet setup
Absorption

Absorption

0.6

0.4

0.2

0.0
0.1

0.2

0.3

0.4

Frequency (THz)

(b)
Figure 3.15 Measured frequency response (a) and absorption (b) of a 0.2 THz cross absorber
using waveguide measurement.
59

Figure 3.16 shows the measured result obtained for a 0.2 THz circle absorber using the
waveguide test setup. The waveguide measurement results of a 0.2 THz circle absorber are
significantly similar to the simulation results. The peak absorption was measured as 82.2 % at
0.2 THz. The effective usable FWHM was characterized to be 25.2 % (bandwidth: 0.182 to
0.232 THz) as depicted in Figure 3.16 (b). As studied above, it was demonstrated that the single
layer absorber provides strong absorption over a wide bandwidth. Further improvements can be
made by double layer stacking. Stacking of a resonant structure leads to wider bandwidth and
also stronger absorption. Design and experimentation were carried out for a hybrid combination
of two 0.2 THz and 0.4 THz absorber layers. This was fabricated using two layers of LCP
dielectrics and pressed together. There is a common solid ground plane present on the back of
these structures.
Figure 3.17 shows the response obtained from a hybrid double stack of absorbers
composed of a 0.2 THz cross absorber with a 0.4 THz cross absorber. Here, the 0.4 THz absorber
is touching the dielectric waveguide and the 0.2 THz design is between the 0.4 THz layer and the
metal backing layer. As expected, this configuration produces two usable absorption bands
(Figure 3.17 (b)). The peak absorptions occur at or around the frequencies of interest (0.225 THz
and 0.45 THz). To be specific, the peak absorptions of dual bands, 95.3 % and 84.3 % were
measured, respectively. Moreover, the FWHM of each band is 61.4% and 21.8%. It is clear that
double stacks of absorbers are proof that multi-band and ultra-wideband absorbing structure can
be fabricated using this approach. This measurement shows that absorption bands can easily be
tailored for different applications. In case of noise suppression, extreme bandwidth with high
absorption is important.

60

20

Measured spectra of a 0.2 THz Circle absorber
on Frequency domain
waveguide
single 0.2THz circle absorber

Magnitude (a.u.)

15

10

5

0
0.1

0.2

0.3

0.4

Frequency (THz)

(a)
Absorption of a 0.2 THz Circle absorber
using Waveguide measuremet setup
Absorption

0.8

Absorption

0.6

0.4

0.2

0.0
0.1

0.2

0.3

0.4

Frequency (THz)

(b)
Figure 3.16 Measured response of a 0.2 THz double layer structure: (a) reference and sample
signal, (b) the absorption.
61

Measured frequency spectra of stacked 0.2/0.4 THz cross absorbers
using Waveguide measurement setup

12

waveguide
stacked 0.2/0.4THz cross absorber

Magnitude (a.u.)

10
8
6
4
2
0
0.2

0.3

0.4

0.5

Frequency (THz)

(a)

1.0

Absorption of stacked 0.2/0.4 THz Cross absorbers
using Waveguide measuremet setup
Absorption

Absorption

0.8

0.6

0.4

0.2

0.0
0.1

0.2

0.3

0.4

0.5

Frequency (THz)

(b)
Figure 3.17 Measured response of 0.2/0.4 THz cross absorber double layer structure: (a)
Measured frequency response, (b) the absorption.
62

3. 7. Conclusions
In this chapter, sets of metamaterial thin-film THz absorbers have been studied through
EM analysis, design and fabrication. Designs having strong absorption and ultra-wide bandwidth
have been demonstrated. The proposed approach takes advantage of skin-effect losses in order to
make high resistivity and wide band metamaterial absorbers. Further increases in bandwidth
were demonstrated through multi-stacking of metamaterial absorbers. The results of proposed
metamaterial absorbers clearly demonstrate a novel approach to achieve ultra-wide band
absorption with strong absorption. THz Absorbers having large FWHM can easily be designed
using the proposed approach. The absorbers were tested using two types of measurement setups
emulating electronic package environments. These absorbers will find applications in electronics
packaging to dampen stray signals inside a package cavity. The proposed designs will also find
applications in sensing and imaging applications.

63

CHAPTER 4. THZ POWER METER
A novel power meter operating over a frequency range of 0.1 to 0.2 THz is presented in
this chapter. The power meter utilizes a thin-film metamaterial inspired absorbing layer. This
absorber has the desired characteristics such as ultra-wide bandwidth and strong absorption. In a
bolometer-like measurement setup, it functions as a wideband high-responsivity THz detector.
The absorbing layer consists of a low loss thin dielectric membrane onto which a twodimensionally periodic, thin metallic layers is patterned. The measured absorption coefficient is
above 0.75 at 220 GHz range.

4. 1. Designs and simulations
For the first design, a cross type and Minkowski fractal resonant structure were used
and details of these were discussed in earlier chapter. In order to simulate these structures, the
Floquet theorem simulation method has been used same as chapter 2. The finite-element method
(FEM) by commercially available software Ansoft HFSS® is performed to produce the
resonance characteristics and modal field distributions of the proposed structures. The cross type
structure is initially studied as planar periodic arrays. In simulation, the metallic element is
defined as nickel with 5,000 Å of thickness. The elements of cross type absorber is considered on
a 50 μm thick liquid crystal polymer, the length of long arm is 400μm, its short arm is 70 μm,
and the gap between the unit-cell is 225 μm.
.

64

(a) Cross type

(b) Minkowski fractal type

Figure 4.1 The layouts and EM field distribution of the bolometers with meta-inspired
absorbers.
Figure 4.1 shows the simulated results of a Minkowski structure that is designed to
operate at 0.2 THz. The frequency of strong absorption is approximately at 0.21 THz. At this
frequency, 9 dB of absorption is attained. It is clear that the absolute density of absorption is
higher than 80 % absorption.

From simulations, the half power absorbing bandwidth is

approximately 60 GHz (between 180 and 240 GHz). The bottom of Figure 4.2 shows simulated
E and H-field distributions for the Minkowski structure as designed for this work. It is clear that
strong E-field appears at the edge of structure and the EM wave penetrates the space between the
metallic layers.

65

Figure 4.2 Simulation results of a 0.1/0.2 THz Minkowski absorber with various incident angles.

66

Figure 4.2 (cont’d)

67

Figure 4.2 (cont’d)

400

Frequency (GHz)

350

TE

300
250
200
150
100
50
0

30

60

90
 (degree)

68

120

150

180

4. 2. Fabrication
The terahertz power meters were fabricated using a conventional micro-fabrication
method. These power meters consist of two parts – a bolometer and a cavity. The bolometer is
composed of 3-layers, Ni/Ti/Cu; 300, 20, and 300 nm, respectively. This structure was deposited
on liquid crystal polymer (LCP) with the thickness of 50 µm. Figure 4.3 shows the fabrication
process using conventional microfabrication method and photographs of bolometers composed
of the metamaterial-inspried absorber structures. Two variations of the metamaterial unit cell
were designed, a regular cross and a Minkowski fractal.

Figure 4.3 Fabrication process of absorbers designed by cross and Minkowski structures.

69

4. 3. Experimental setup and measurement results
The transmission measurement setup has been used to characterize the 0.1 and 0.2 THz
absorbers in this work. Figure 4.4 shows the measured absorptance of proposed bolometers in
frequency domain. Figure 4.4 (a) shows a maximum absorptance of 0.48 and strong overall
absorptance between 0.065 and 0.18 THz. Figure 4.4 (b) demonstrates the response obtained
from a Minkowski fractal type bolometer designed to have a response at two adjacent bands.
This configuration produces a second absorption band (0.05 THz up to 0.25 THz). The peak
absorptions occur at or around the frequencies of interest (0.05 THz and 0.22 THz). This
absorber is proof that multi-band and ultra-wideband absorbing structures can be fabricated and
applied to wide-band power detecting on specific frequency ranges that is willing to measure the
incident power density.
In order to use it as a bolometer, the resistivity temperature coefficient is normally an
important characteristic. Actually, the electrical resistance of a typical metal increases linearly
with rising temperature. This is because the surface resistance changes depending on the amount
of incident power being converted to heat and in turn change in the resistance of the film. The
measurement setup of resistivity temperature coefficient is depicted in Figure 4.5. The thickness
of metal deposition is Ni/Ti/Cu; 300, 20, and 300 nm.

70

Frequency Domain
0.5

Absorptance

0.4

0.3

0.2

0.1

0
0.05

0.1

0.15

0.2

0.25

Frequency(THz)
(a). Cross-type bolometer

Frequency Domain
0.8
0.7

Absorptance

0.6
0.5
0.4
0.3
0.2
0.1

0.05

0.1

0.15

0.2

Frequency(THz)
(b). Minkowski fractal type bolometer
Figure 4.4 Measured absorptance of proposed bolometers.

71

0.25

0.3

Figure 4.5 shows the measured temperature dependence of Ni film as a function of
applied temperature measured by placing the sample in a vacuum oven and regulating the
temperature. The temperature of the oven was slowly increased and the allowed to settle down
before taking the measurements. A thermometer placed in the oven was used to measure the
oven temperature. As the temperature increases, the resistance of the film increases linearly.
Table 4.1 summarizes the measured surface resistance of bolometers using a four point probe
measurement technique [45, 46].

Table 4.1 Measured resistance of proposed bolometers
Type of bolometer

Resistance(Ω)

Cross(positive)

440 Ω

Cross(negative)

15.2 Ω

Minkowski fractal

52.4 Ω

The bolometer structures were measured using a high frequency source, Backward
Wave Oscillator (BWO 75~110 GHz) at 110 GHz. Figure 4.5 shows the layout of the
measurement fixture. The periodic structure absorbs the incident radiation which increases the
surface temperature of the metallization and increases the resistance of the aggregate structure.
This variation in the resistance is measured and used to characterize the performance of the
bolometers. In Figure 4.6 (a), the bolometer is located on a thick Styrofoam layer to reduce
reflection and mitigate frequency change. Figure 4.6 (b) shows the equivalent circuit of testing
schematic including the styrofoam used in decreasing the reflectance [47].

72

1.08
y = 0.002698295x + 0.940021705

1.07
1.06

R/R0

1.05
1.04
1.03
1.02
1.01
1
0.99
23

28

33

38
Temperature(°C)

43

48

Figure 4.5 Measurement setup and measured result of resistivity temperature coefficient of
nickel (Ni).

73

(a)

(b)
Figure 4.6 Quasi-optical measuring schematics of the bolometers using a high frequency source.

74

Also, we measure two different setups, one is a transmission mode which is composed
of the bolometer and a styrofoam, thus the receiving power is able to measure. Another
measurement is the reflection mode which comprises of the bolometer, a Styrofoam layer, and
the cavity. Using this configuration, the density of incident wave arrived at the bolometer can be
enhanced due to multi reflection from the cavity [47].
Figure 4.7 demonstrates the resistance variation of the bolometers, as a function of
incident power. In transmission mode measurement, the variation slope of the Minkowski
bolometer is the steepest among the proposed bolometers. In reflecting mode, the positive cross
bolometer is more sensitive than the Minkowski bolometer.

(a) Transmission mode
Figure 4.7 Measured results of normalized resistance variation of the bolometer, (a)
Transmission mode, and (b) Reflection mode.
75

Figure 4.7 (cont’d)

(b) Reflecting mode

76

4. 4. Conclusions
This chapter demonstrates the design and fabrication of a new class of power meter
using metamaterial-inspired thin-film absorbers operating at millimeter and sub-millimeter
wavelengths. For a cross-type bolometer, the absorption bandwidth results in approximately 0.12
THz. A Minkowski bolometer shows dual band that can be utilized to measure the power density
in a multi-band system. In quasi-optical measurement results, all of proposed bolometers have
shown to have high sensitivity due to the characteristic of high surface resistivity by skin-depth
effect and absorbing of metamaterial-inspired structures. Compared to conventional power
meters wherein the response is achieved purely due to material characteristics, these designs also
utilize field enhancement due to resonance of the metallization to achieve enhanced sensitivity.
In other words, resonant structures can be designed to directly absorb the incident radiation
without resorting to antenna coupling or use of large area structure with back reflectors. The
proposed power meters also have several advantages such as low cost, simple fabrication, and
easy integration with other monolithic circuits.

77

CHAPTER 5. TERAHERTZ SURFACE PLASMONICS BULLS-EYE
ANTENNA
5. 1. Theoretical background
Recently, plasmonic applications play a significant role in THz regime. THz absorbers
introduced and discussed in previous sections basically work as the mechanism related to the
THz surface plasmons at metal and dielectrics in frequency selective surfaces. In this chapter, the
concept of plasmonic phenomenon will be introduced and the THz antenna, bulls-eyes type,
demonstrated and characterized in order to focus the beam to improve the transmitting power to
the bolometer that has been applicable to high-resolution imaging of THz spectrum.
Surface plasmon is one of physical phenomena. It is the free electrons oscillation at the
boundary between a conducting and a dielectric material. Electromagnetic forces is entrapped at
the boundary go with the wavering surface charge density. There is strong coupling between the
electromagnetic wave and a charge density oscillation that is called a surface plasmon polariton
[48]. Because of the surface charges, the electric fields have components perpendicular to the
boundary, while the components of magnetic fields propagate transverse.
When the surface Plasmon travels in the x direction, the electric and magnetic fields in
the two materials are described as

5 – (1)
5 − (2)

78

5 – (3)
5 − (4)
where Ed and Hd are in the dielectric layer (z > 0), and Em and Hm in the conductor (z
< 0). From these equations, the surface wave is able to be expressed by the wave-numbers kx, kd,
and km for an operating frequency ω. The dispersion relations are derived by the wave equation,

: wave equation

5 – (5)

5 − (6)

where εd and εm are the dielectric constants of the dielectric material and metal (all of
material are considered nonmagnetic). Merging these two equations, it can be gotten the relation
of the ratio kd/km expressed in terms of kx and ω:

5 – (7)

Applying Maxwell’s equation, it can be obtained the relations between Ex and Hy:
5 − (8)

79

5 – (9)

Ex and Hy , which are parallel with the surface, propagate at the boundary continuously.
The boundary conditions are set as
5 – (10)

Substituting Equation 5-(10) into Equation 5-(5) and (6), the dispersion relation of
surface plasmons can be derived,
5 – (11)

5. 2. Metallic Bulls-eye antenna
5.2.1 Design and simulation
It is well known that the transmission of electromagnetic waves can be enhanced when
passing through subwavelength apertures at optical frequencies [49-55]. This is due to the
constructive interference of resonating plasmonic modes [50]. These transmitted modes can be
modified both spatially and temporally through hole arrays and periodic structures.

Such

systems produce enhanced, narrow-beam transmission while requiring only single wavelength
geometric sizes [49, 53-55]. These apertures can be combined with gratings to couple incident
electromagnetic radiation and enhance their transmission properties even further.
Numerous such plasmonic grating designs have been proposed, many of which are
easily translated to the terahertz frequency spectrum [59]. In fact, several subwavelength aperture
works have been published operating at terahertz frequencies [59-62]. The majority of these
80

systems involve surrounding a central subwavelength aperture with concentric grooves, forming
a spoof-surface plasmon antenna or bulls eye structure.

The grooves redirect incident

electromagnetic radiation toward the subwavelength aperture by exciting surface waves in the
form of surface plasmons [55]. These waves are then transmitted through the aperture at
enhanced levels with subwavelength focusing [49, 51 - 55]. This signal can be further shaped
through the inclusion of a reciprocal grating on the backside of the system.

The major

advantages of the bulls eye system selected include its simpler fabrication, its planar structure, its
relatively polarization-independent performance, and its behavior at terahertz frequencies
spectrum is well known [60 -62].
Recently, this bulls eye structure was paired with a microbolometer for the sensing of
terahertz electromagnetic waves [63]. Here the vanadium bolometer sensor was replaced with a
metamaterial-inspired structure tuned to the plasmonic frequency of the bulls-eye [44, 64]. This
structure is simpler to fabricate and provides high sensitivity. The incoming THz radiation is
focused onto the bolometer elements using the surface plasmon antenna. The THz radiation
absorbed by the metamaterial bolometer element is converted into heat which in turn heats the
bolometer film. The resistance of the bolometer film is sensitive to temperature and its change in
resistance is proportional to the power of the incoming signal. A THz image is formed by
measuring the change in resistance at each pixel forms its reference. This coupled system is
expected to not only capture a large percentage of the incident electromagnetic radiation, but also
efficiently convert this energy to electrical signals at the desired frequency band.
The system was numerically simulated in stages using both finite difference and finite
element electromagnetic models.

First, the bulls-eye structure was characterized for its

transmission properties. Then, the metamaterial bolometer was simulated for its frequency

81

response. Three-dimensional finite element simulations were performed using the commercial
software package, High Frequency Simulation Software (HFSS) at the nominal operating
frequency of 100 GHz. Plane wave excitation from the top side of the bulls-eye structure was
applied. Radiation boundary conditions were applied to all other domain edges. The resulting
electric field plots are provided below in Figure 5.1. The wave travels along the grating and
groove and travels through the centered aperture, finally converging to a small focal point.

Figure 5.1 HFSS simulation result of Electromagnetic field distribution in the bulls eye structure.
This system was also numerically investigated using a two-dimensional finitedifference time domain (FDTD) simulation. Here, a consistent grid spacing of 20 μm was
applied in both directions, with a Gaussian weighted exciting wave in the frequency domain.
The metal structure was assumed to be a perfect electric conductor. Ten perfectly matched
layers surrounded the simulation domain boundaries. Plots of the temporal and frequency
domain transmitted transverse electric fields are presented in Figure 5.2. It is clear from these

82

images that plasmonic modes reside above the bulls-eye structure and travel toward the aperture,
while the transmitted field is tightly focused.

Figure 5.2 Simulated transverse electric field by FDTD.

83

Figure 5.3 shows the response of the plasmonic structure in both frequency and time
domains. The temporal signal demonstrates the transmitted signal that is strongly dependent on
the number of groove periods present. This highly periodic temporal signal leads to an enhanced
frequency response at the designed frequency of slightly less than 100GHz. Moreover, the
relative frequency response shows an increased response.

84

(a)

(b)

Figure 5.3 The simulation results of the bulls eye structure in time and frequency domain using
FDTD. (a) Time-domain transmitted signal, (b) frequency domain transmitted signal, (c)
transmitted signal (dB), and (d) incident Gaussian signal.

85

Figure 5.3 (cont’d)

(c)

(d)

86

5.2.2 Fabrication
The surface Plasmon-assisted terahertz image array is composed of two-parts; the bulls
eye plasmon-assisted antenna, and the metamaterial-inspired bolometer. The fabrication of these
components will be described separately.
5.2.2.1. Layout of the surface plasmon-assisted THz imaging array
Figure 5.4 shows the schematic of the bulls-eye plasmon-assisted antenna. The period
and width (w1 and w2) of the grating is designed to match a half wavelength at the operating
frequency. The diameter (D) of the center aperture was set as three quarters of this wavelength.
The detailed physical dimensions are summarized in Table 5. 1.

Figure 5.4 Schematics of the bulls-eye spoof plasmon antenna.

87

Table 5.1 Physical dimensions of the bulls-eye plasmonic-assited antenna
Bulls-eye antenna structure
100 GHz

300 GHz

Width (W)

1.5 mm

0.5 mm

Diameter (D) of the aperture

2.25 mm

0.75 mm

Depth of grooves (h) and the
aperture (T)

h= 300 μm

h= 100μm

T= 450 μm

T= 150μm

5.2.2.2. Metamaterial-inspired bolometer
The metamaterial-inspired bolometer is based on design in [44, 64].

Figure 5.5

illustrates the layout and the dimensions of the metamaterial-inspired bolometer designed for
operation at THz frequencies. The Minkowski fractal type pattern has been known to work at
multi-band and wide band frequency ranges as a power meter [61-64].

Figure 5.5 Schematic of a Minkowski fractal that forms a unit cell of a bolometer (Shown are
the field strengths at resonance frequency).

88

5.2.2.3. Fabrication of the bulls-eye plasmon-assisted antenna
The bulls-eye plasmon-assisted antenna was fabricated using conventional metaletching micro-fabrication process. A high purity copper plate was chosen as the base substrate
for the bulls-eye structure fabrication. After photoresist patterning the Cu-plate was wet etched.
Figure 5.6 shows photograph of a fabricated structure.

Figure 5.6 Photograph of the Bullseye Plasmonic-assisted antenna (100 GHz design).

5.2.2.4. Fabrication of the metamaterial-inspired bolometer
The metamaterial-inspired bolometers were fabricated using conventional microfabrication methods. The bolometer is composed of 3-layers, Ni/Ti/Cu :: 300/20/300 nm,
respectively. Titanium (Ti) enhances adhesion, nickel (Ni) is the absorbing and the resistance
layer, and Cu was used on the pad regions. This structure was deposited on a 50 µm thick liquid

89

crystal polymer (LCP) substrate. Details of bolometer detector are discussed in [64]. Figure 5.7
shows photograph of a fabricated bolometer composed of an array of Minkowsi structures.

Figure 5.7 Photograph of a Minkowski fractal type metamaterial bolometer.
5.2.3 Experimental setup
In order to characterize the surface plasmonic-assisted image array, time domain and
frequency domain measurement systems were both used. A time domain system, Picometrix Tray 2000 THz time domain system (TDS) fitted with both a receiver and transmitter was used for
the measurements of the bolometer and the bulls-eye plasmonic-assisted antenna. This
measurement system provides a frequency resolution of approximately 2 GHz [44]. The
transmitter radiates a collimated electromagnetic pulse beam in the direction of the receiver. The
time domain setup excites an electromagnetic wave impinging normally upon the device under
test. Using this setup, the characteristics of the bulls eye plasmonic-assisted antennas and

90

metamaterial-inspired bolometers were measured. A block diagram of the time domain
measurement setup is provided in Figure 5.8.

Figure 5.8 . Block diagram and photograph of the time domain measurement setup.
The frequency domain measurement setup is comprised of a spectrum analyzer and a
W-band (75 – 110 GHz) backward wave oscillator (BWO). The measurement setup used is
similar to a setup used in far-field test of the antenna radiation pattern [65]. In order to generate a
plane wave, the transmitter and receiver were positioned within the far-filed range associated
with a W-band reference horn antenna (HPBW=12°). Furthermore, to extract the properties of
the plasmonic-assisted pattern, a reference signal was obtained from a pin hole of similar
diameter as the bulls-eye aperture and the same metal thickness (T). Based on the measurement
91

of the bulls-eye antenna and pin-hole, the performance of plasmonic assisted pattern may be
estimated. Figure 5.9 shows the experimental setup for the frequency domain measurement
system. A 3-axis manipulator was used to align the boresight of the bulls-eye plasmonic-assisted
antenna.

Figure 5.9 Configurations of the frequency domain measurement setup.

92

5.2.4 Measured results
5.2.4.1. Characterization of the bulls-eye plasmonic antenna
To verify the performance of the bullseye plasmonic-assisted antenna, the antenna was
measured in both time and frequency domains. Figure 5.10 shows the characteristics of 100 GHz
bulls-eye antenna in the time domain. The measured result demonstrates that the maximum
transmitted frequency occurs at approximately 95GHz. Figure 5.11 shows the measured results
of the reference (no antenna) and transmission antenna and pin-hole at 96 GHz. In the frequency
domain, the received signal from the bulls-eye plasmonic-assisted antenna is measured -45.3
dBm and that of the pin-hole is -57.3 dBm. Background noise floor is roughly -71 dBm.
These measurement results were then converted to the signal to noise ratio (SNR). This
was done by subtracting the SNR of the pin-hole from the SNR of that shown in Figure 5.12. The
SNR of the bulls eye antenna is significantly higher than the pin-hole structure (approximately
16 times at 96 GHz).
7
Pinhole
Bulls eys Plasmonic

Transmission Magnitude (a.u.)

6

5

4
3

2

1

0
0.5

0.6

0.7

0.8

0.9

1
Frequency (Hz)

1.1

1.2

1.3

1.4

1.5
11

x 10

Figure 5.10 Measured results of the bulls-eye plasmonic-assisted antenna using the THz time
domain measurement system.

93

(a)

(b)
Figure 5.11 (a) Transmitted signal through the bulls-eye plasmonic-assisted antenna
characterizing at 96 GHz (-45.3 dBm); (b) transmitted signal through a reference pin-hole (-57.3
dBm). Noise floor: -71 dBm at BW: 1 GHz.

94

12
Deviation

10

Gain (dB)

8
6
4
2
0
-2
90

92

94

96

98

100

Frequency (GHz)
Figure 5.12 Measured signal to noise ratio (SNR) of the bulls eye structure relative to the pin
hole.

95

5.2.4.2. Measurement results of the proposed 100 GHz imaging array
In order to further characterize the bull’s eye structure, the structure was coupled to a
bolometer structure. This allows in direct measurement of the transmitted power. Figure 5.13
illustrates the test setup. The bolometer structure is closely placed (~ 0.5 mm) near the aperture.
A W-band antenna coupled harmonic mixer was used to calibrate the incident power from the
BWO. This was carried out prior to the placement of bull’s eye and bolometer structures. Figure
5 .14 shows the measured change in resistance of the bolometer as a function of frequency of the
incident signal on the bull’s eye structure. The measured results show a maximum detection at
96GHz. This measurement shows that the bull’s eye structure can be used as a narrow band
antenna. Also, it can be closely coupled to the bolometer structure while avoiding thermally
loading the bolometer. The bolometer is compatible with planar fabrication and thus can be
utilized in the fabrication of bolometer based THz imaging array.

Figure 5.13 Diagram of the measurement setup for Image array.

96

0.07

Resistance deviation (ohm)

0.06
0.05
0.04
0.03
0.02
0.01
0.00
90

92

94

96

98

Frequency (GHz)

Figure 5.14 Measured resistance deviation of the bolometer detector element.

97

100

5. 3. Plastic Bulls-eye antenna
5.3.1 Design and simulation
In order to analyze and verify the behavior of the proposed bulls eye structure,
numerical simulations were carried out using both the two dimensional finite difference time
domain (FDTD) method and a commercial finite element method (FEM) software package
(Ansoft® HFSS). Two plastic materials were analyzed in the design of antenna element; acrylic
(εr=2.9) and high density polyethylene (HDPE, εr=2.37). The metal layer was approximated as a
perfect electric conductor in the FDTD simulation, and assumed to be copper in the HFSS
simulations. The FDTD simulation results for the transverse electric (TE) fields and transmission
frequency response are provided in Figure 5.15. Here, it is clear the surface plasmons reside on
the metallic groove surface located inside the plastic material. The electromagnetic wave is
focused toward the center aperture by the SPs. In the frequency response plot, the operating
resonance frequency is approximately 100 GHz.
FEM simulations were also carried out using the commercial EM tool, Ansoft® HFSS
v.13 for the proposed antenna structure. Plane wave excitation from the top side of the bulls eye
structure was applied. Radiation boundary conditions were applied to all other domain faces. The
resulting wave traveled along the grating and groove through the center aperture, finally
converging to a small focal point. In this simulation, the plastic material was assumed as HDPE.
This simulation result is shown in Figure 5.17. Good agreement between the two simulations is
observed, with the majority of the electromagnetic energy being concentrated within the plastic
material at the groove surface. A significant portion of the energy also surrounds the center subwavelength aperture.

98

(a)

Magnitude
(Hz)
(b)
Figure 5.15 FDTD simulation results of a plastic (acrylic) bulls-eye antenna: (a) E-field
distribution, (b) Transmitted signals.

99

Figure 5.16 FEM (HFSS) simulation result of E-field distribution in the plastic bulls eye antenna.

5.3.2 Fabrication
A 3-D plastic printer was utilized in the fabrication of the proposed plasmon structure.
The printer provides high resolution printing (resolution of ~100m). The back side of the
structure is metalized and this was achieved by sputter depositing thin titanium (Ti) and copper
(Cu) layers. Titanium improves the adhesion of the Cu to the acrylic layer. Figure 5.17 shows the
pictures of the fabricated bulls-eye structure.

Figure 5.17 Photographs of a fabricated plastic bulls-eye antenna.

100

5.3.3 Measurement setup
In order to characterize the fabricated bulls eye antenna, the quasi-optic far-field
measurement setup was used as shown in Figure 5.9. Also, the fabricated antenna is placed in the
middle of the distance between a source (BWO) and a receiver (W-band harmonic mixer) as
same measurement setup as the metallic bulls eye antenna. Figure 5.18 (a) demonstrates
photographs to set up the measurement. It is to calculate the performance of the plastic antenna
with trenches (leading to the plasmonic phenomenon), which a reference signal was acquired
from a pin hole of same diameter as the bulls eye aperture. Using this measurement setup, we
may investigate and verify the idea that how the plastic layer works to trap the EM wave and
enhance the transferring power to the hole traveled through the plasmonic patterns. Also, the
relative gain can be obtained to compare the received power between a pin hole and the plastic
bulls eye antenna. The area of the metallic pinhole and a plastic antenna covers the receiving
horn antenna entirely. This configuration helps avoid the interfering signals from the
environmental sources. The diameter of aperture size of all samples (a metallic pinhole, a plastic
pinhole, and a dielectric bulls eye antenna) is set as 1.5 mm (depicted in Figure 5.18 (b)).
All samples have been measured to sweep operating frequencies from 75 to 110 GHz
with 1 GHz step. To tune up the operating frequency, the indicator of BWO must set as the
specific value for operating frequency. The detail set values of a BWO have been summarized in
Appendix B. Before measuring for samples, we need to calibrate and analyze the source (BWO
and W-band horn antenna). The estimated gain and effective area of a horn antenna is 15 dBi
2

(referring to the gain of a standard pyramid horn) and 456 mm (W × H=24mm × 19mm). The
rated power of a BWO is measured 240 μW at 100 GHz. Thus, the estimated effective isotropic
rated power (EIRP) is computed to be 8.8 dBm (power from the flange out of a horn antenna).
101

(a)
metallic pinhole

plastic pinhole

(b)
Figure 5.18 Photographs of measurement setup of a plastic bulls-eye antenna.

5.3.4 Measured results
As discussed above, in order to measure the characteristics of a plastic bulls-eye
antenna, firstly, the metallic pinhole and a plastic pinhole have been measure to verify the
trapping effect of a dielectric layer (ABS). Secondly, in order to evaluate the plasmonic pattern
on the plastic layer, the ABS pinhole and the plastic bulls-eye antenna have been tested. As
results of this measurement, the bulls eye antenna can be characterized how large the gain can be
achieved comparing to the ABS pinhole. Figure 5.19 demonstrates the received power of a
metallic pinhole and ABS pinhole that transmitted through the harmonic mixer of a spectrum
analyzer. Over all of frequencies, the plastic pinhole shows higher (3~7dB) transmitting
characteristic than a metallic pinhole. To be specific, the received power of a metallic pinhole is
102

close to the noise level of a channel, -80 ~ -82 dBm. At 89 GHz, both of samples show similar
received power, it is because the power from a BWO drops rapidly at this frequency. Around 97
GHz to 102 GHz, metallic pinhole has been measured higher amount of received power
comparison to rest of frequencies range, because of the aperture size of a pinhole (a half wave
length of 100 GHz).
Metal pinhole
ABS pinhole

Received power (dBm)

-72
-74
-76
-78
-80
-82
70

75

80

85

90

95 100 105 110

Frequency (GHz)
Figure 5.19 Measured received power of a metallic pinhole and plastic (ABS) pinhole.

Figure 5.20 shows the received power obtained from the ABS pinhole and a plastic
bulls-eye antenna composed of plasmonic trenched pattern. Likewise the previous measurement,
both of plastic samples has been measured from 75 GHz to 110 GHz. As expected, the received
power of the antenna having a trenched pattern shows higher than a pinhole. In measured results,
we can figure out that there is the significant usable band, from 75 GHz to 87 GHz, the

103

bandwidth is 12 GHz. To convert to the relative gain, the plastic bulls-eye antenna is computed
to be 15 dB (maximum gain at 83 GHz) compared to a pinhole.

Received power (dBm)

-60

ABS pinhole
Bulls eye

-65
-70
-75
-80
-85
70

75

80

85

90

95 100 105 110

Frequency (GHz)

Figure 5.20 Measured received power of a plastic (ABS) pinhole and plastic bulls-eye antenna.

104

5. 4. Conclusions
This research provides details of simulation, fabrication, and experimental analysis of a
spoof-plasmon assisted antenna coupled with a metamaterial-inspired bolometer for THz
imaging. Enhanced power transmission and detection was reported at 96 GHz for the bulls eye
antenna compared to a pin-hole structure. Due to its low cost and ease of fabrication, this system
is expected to be easily scaled to large-scale imaging arrays. The proposed approach provides an
approach to integrate a bolometer structure and the antenna element while avoiding thermal
loading. Furthermore, bolometers that are subwavelength in size can be used. Small bolometer
size is necessary in order to improve signal to noise ratio.
 In this chapter, a bulls eye antenna utilizing surface plasmonic enhanced transmission has
been designed, analyzed and fabricated on a plastic substrate.
 The combination of surface plasmons and plastic materials help enhance the transmission
power compared to similar bare metal bulls-eye antennas. Furthermore, as use the plastic
material to fabricate the antenna, it can help improve the transmitting power as well.
 The proposed antenna design may be integrated with a bolometer to produce a power
detector. This structure is simple to fabricate and is wafer-level processing compatible.
Also, this structure can be adopted for circuit design at optical frequencies.

105

CHAPTER 6. DIELECTRIC MICRO LENS ARRAY FOR THZ FOCAL
PLANE
6. 1. Theory of Lens
Several lens designs have been presented in literature: hemispherical, hyper/hypohemispherical, and extended hemispherical lenses [66-73]. The extended hemispherical lens is
attractive as it offers higher directivity and can be used in imaging array systems [67]. Other
advantages include higher collimating beam or sharper radiation pattern, and are interoperable
with Gaussian beam systems [66]. In this paper, we studied two types of lenses: (i) extended
hemispherical and (ii) hypo-hemispherical lenses. Schematics of the dielectric lenses are shown
in Figure 6.1. Large diameter (≥ 10λ0) lenses have been studied in great detail to achieve high
directivity [70-72]. However, in THz focal plane systems designs, there is a need to meet the
higher density integration challenge and thus smaller diameter lenses are needed.

(a)
(b)
Figure 6.1 The schematics of proposed dielectric lenses: (a) extended hemispherical, (b) hypohemispherical.

106

In order to design the extended hemispherical lens, the hemispherical radius and
extended length must be calculated. For this calculation, the ellipse equation is utilized [68]:
2

2

 x
 y
    1
a
b
Where, the foci are at ± c and

6 − (1)

c  b2  a2

The eccentricity (e) of the ellipse can be analyzed with the refractive index, n, that is
characterized by the optics:

b2  a 2 1
e

b
n
b

a
1
1 2
n

,

b
c
n

6 − (2)

6 − (3)

For analyzing electrical path lengths of dielectric lenses, the following Fermat’s
principle was used [72],

R

0

R



F

0



R cos   F

d

(n  1) F
n cos   1
107

6 − (4)

6 − (5)

where R is the radius from radiating source to the lens surface, F is the focal point, and
λo and λd are wavelength of free-space and guided wavelength in a dielectric material,
respectively. Based on these equations, the dielectric ellipsoidal structure can be calculated.
However, practically a perfect elliptical lens can be hardly produced. Thus, the extended
hemisphere lens where the feeding or bolometers are slightly offset from the center of a designed
hemisphere can be analyzed by a desired dielectric material extension of thickness, Lext. The
optimal ratio of the radius of a designed hemisphere and the length of extended part was referred
Lext/R=0.39 that is suitable to the ellipse with eccentricity (ε=1/η=0.65) for HDPE, η=1.543 [74].

6. 2. Simulation
THz lenses are to be made out of dielectric materials, specifically, high density
polyethylene (HDPE, εr=2.35), and Acrylonitrile Butadiene Styrene (ABS, εr=2.79). These
materials are cost effective, easy to produce, and show low dielectric loss in the THz frequency
range [43, 75]. Dielectric properties of HDPE in the THz frequency range are presented in [75].
In this paper ABS material is characterized using a method presented in [43]. The measured
results will be presented in detail in chapter 6.5. Physical dimensions, such as the diameter and
the extended length of the extended hemispherical dielectric lens, are calculated by the equations
explained above. Table 6.1 details the key physical dimensions.
Ansoft High Frequency Simulation Software (HFSS®) was used to simulate the
proposed THz lens design based on three-dimensional Finite Element Method (FEM). In order to
analyze the proposed lenses, simulations of the lenses were divided into two categories: (i) field
simulations of electromagnetic waves and (ii) radiation pattern simulations such as near-field and
far-field which are typically used in antenna design analysis.
108

Table 6.1 Physical dimensions of extended hemispherical lenses

D = 1.6 × λ0

D = 5 × λ0

Hypo

HDPE Lens(εr =2.35)

Acrylic Lens(εr=2.79)

R1 = 0.80mm

R1 = 0.80mm

L1 = 1.02mm

L1 = 0.93mm

R1 =2.50mm

R1 = 2.50mm

L1 = 3.19 mm

L1 = 2.93 mm

R2 = 1.25 mm

R2 = 1.25 mm

L2 = 0.65mm

L2 = 0.60mm

Figure 6.2 shows the E-field distribution for two extended hemispherical lenses (D =
1.6 × λ0, 5 × λ0) and a hypo-hemispherical lens. In the field simulation, plane waves were
simulated starting from the bottom side of dielectric lens to analyze the E-field distribution.
While passing through the lens structure, the electromagnetic wave focuses on a point, the focal
point, which shows a high intensity of the traveling wave. Figure 6.2 (a) shows that the
collimating beam is broad compared to the lens diameter and the focal point is located 1.33 mm
away from the plane wave source. Figure 6.2 (b) and (c) illustrate the simulated E-field
distribution of a large diameter (D = 5 × λ0) lens and a hypo-hemispherical lens. In both of the
simulated results, the main beam is narrower and enhances the transmission intensity more than a
smaller diameter lens. This is predicted by theory and results from an increased effective lens
diameter [64].

109

(a)
(b)
(c)
Figure 6.2 Results of incident wave field analysis for extended hemispherical lenses (a) D = 1.6
× λ0, (b) D = 5 × λ0, and (c) hypo-hemispherical lens.
For the near field simulations, the boundary is located on the focal point of each lens
where the power intensity of travelling wave is concentrated. Figure 6.3 shows the near field
simulation results of two hemispherical lenses (D = 1.6 × λ0 and 5 × λ0) and a hypohemispherical lens. All simulated radiating beam shapes clearly demonstrate the typical Gaussian
beam distribution. Among the simulation results, a hypo-hemispherical lens has a narrow beam
and low side lobe. However, the focal point of the hypo-hemispherical lens is located 3.61 mm
away from the plane wave source, which is three times further away than the simulated small
diameter hemispherical lens (D = 1.6 × λ0).

110

Near-field radiation

HFSSDesign1

ANSOFT

100.00
Curve Info
dB(NearETotal)_1
Setup1 : LastAdaptive
Freq='300GHz' Phi='0deg'

dB(NearETotal)_1

90.00
80.00
70.00
60.00
50.00
40.00
-180.00

-90.00

0.00
Theta [deg]

90.00

180.00

(a)
XY Plot 2

5lambda

ANSOFT

90.00
Curve Inf o
dB(NearETotal)
Setup1 : LastAdaptive
Freq='300GHz' Phi='0deg'
dB(NearETotal)
Setup1 : LastAdaptive
Freq='300GHz' Phi='90deg'

dB(NearETotal)

70.00

50.00

30.00
-90.00

-45.00

0.00
Theta [deg]

45.00

90.00

(b)
Figure 6.3 Simulated near field (E-field) radiation patterns of extended hemispherical lenses: (a)
D = 1.6 × λ0, (b) D = 5 × λ0, and (c) hypo-hemispherical lens.

111

Figure 6.3 (cont’d)
Near_field_hypo

lens_hypo

ANSOFT

90.00
Curve Info
dB(NearETotal)
Setup1 : LastAdaptive
Freq='200GHz' Phi='0deg'
dB(NearETotal)
Setup1 : LastAdaptive
Freq='200GHz' Phi='90deg'

dB(NearETotal)

80.00

70.00

60.00

50.00
-90.00

-60.00

-30.00

0.00
Theta [deg]

30.00

60.00

90.00

(c)

Figure 6.4 demonstrates the far-field radiation patterns (directivity of E-plane) which
were calculated using HFSS. In this simulation, all of lenses have high directivity (> 15 dB) at
the target frequency. In addition, as the diameter of a given lens increases the directivity
improves. The maximum directivity of a hemispherical lens (D = 5 × λ0) is computed as 23.5 dB
at 300 GHz with HFSS while that of a hemispherical lens (D = 1.6 × λ0) is 16.1 dB at same
operating frequency. It is clear, then, that the large diameter lens has a higher directivity than the
small diameter lens. A hypo-hemispherical lens shows similar performance characteristics to a
large hemispherical lens when both lenses have the same radius (1.6 × λ0).

112

XY Plot 1

5lambda

30.00
Name
m1

X

ANSOFT

Curve Info

Y

0.0000 23.5810

dB(GainTotal)
Setup1 : LastAdaptive
Freq='300GHz' Phi='0deg'

m1

dB(GainTotal)
Setup1 : LastAdaptive
Freq='300GHz' Phi='90deg'

20.00

dB(GainTotal)

10.00

0.00

-10.00

-20.00

-30.00
-90.00

-60.00

-30.00

0.00
Theta [deg]

30.00

60.00

90.00

(a)
XY Plot 3

1_6lambda

ANSOFT

20.00
Curve Info
dB(GainTotal)
Setup1 : Sw eep
Freq='300GHz' Phi='0deg'
dB(GainTotal)
Setup1 : Sw eep
Freq='300GHz' Phi='90deg'

dB(GainTotal)

0.00

-20.00

-40.00
-90.00

-60.00

-30.00

0.00
Theta [deg]

30.00

60.00

90.00

(b)
Figure 6.4 Simulated far field pattern of the two extended hemispherical lenses: (a) D = 5 × λ0,
(b) D = 1.6 × λ0, and (c) hypo-hemispherical lens.
113

Figure 6.4 (cont’d)
Far_field_hypo

lens_hypo

15.00

ANSOFT

Curve Info
dB(GainTotal)
Setup1 : LastAdaptive
Freq='200GHz' Phi='0deg'
dB(GainTotal)
Setup1 : LastAdaptive
Freq='200GHz' Phi='90deg'

dB(GainTotal)

5.00

-5.00

-15.00

-25.00
-90.00

-60.00

-30.00

0.00
Theta [deg]

30.00

60.00

90.00

(c)

The radiated electric field has been demonstrated in Figure 6.5. Electromagnetic waves
have been exited from the bottom port. In order to analyze the field distribution, the wave port
has been set the waveport in HFSS. On a simulated result of a 5×λ0 lens, after transmitting the
lens, the propagating wave was transformed to the plane wave. Similarly, the small diameter (1.6
× λ0) of a lens shows the propagating plane wave.

114

(a)
Figure 6.5 The electric far-field radiation pattern of 5 × λ0 (a) and 1.6 × λ0 (b) lenses.

115

Figure 6.5 (cont’d)

(b)

116

6. 3. Fabrication
Two methods were studied for the fabrication of micro-lens array. First, a 3-D plastic
printer was utilized in the fabrication. The printer provides high resolution printing (resolution up
to 30 m). ABS is used as the dielectric material (detail measured material properties are
presented in chapter 6.5). This fabrication method has several advantages: time-saving
fabrication, good reproducibility, and allows prototype fabrication in a short cycle time. Figure
6.6 shows photographs of fabricated lenses using this approach. The micro molding was used in
a second fabrication process. Details of this process will be presented at the conference. This
method is desirable when mass production of a design is required. Furthermore, this method
allows larger flexibility in use of dielectric materials (thermoplastic materials) as compared to
3D printing. Generally, in order to use the micro molding process, the metallic mold must be
produced. For this, normally, micro-machining or micro fabrication has been used [72]. In this
manuscript, a new cost effective way to manufacture the mold structure using 3D printing and
metal plating was utilized. As a first step, starting with the 3D printed structure, the 3D structure
is coated with thin titanium (Ti) and Cu seed layers for electroplating. Bulk metal plating was
used to build up the thick metal layer (up to mm level). The base 3D plastic structure is dissolved,
using solvents, leaving behind a thick Cu mold structure. This mold structure is mounted on a Cu
plate through soldering and used as the mold for fabrication of HDPE based micro-lens array.

117

100 µm
(a)

200 µm
(b)

200 µm
(c)
Figure 6.6 Photographs of fabricated THz lenses (left: lens array, right: microphotographs of
lens surface).

118

6. 4. Experimental measurement setups
In order to measure the fabricated THz lenses, we used two measurement methods;
near-field and far-field testing. Both measurements utilized the time domain measurement
system (Picometrix T-ray 2000 THZ TDS). This measurement system offers a sampling period
of 0.03941ps [44]. This result in a frequency resolution of approximately 12.5 GHz when
acquiring the Fourier transform of the received time domain signals. The transmitter radiates a
collimated electromagnetic pulse beam in the direction of the receiver.
The block diagram of far-field measurement setup is

demonstrated in Figure 6.7. In

order to apply the plane incident wave, the distance between the transmitter and a THz lens was
2

set as 30 cm (>> 2d /λ) in far-field range [65]. At this distance, the main beam of a transmitter
covers up approximately 2.5 inches diameter on the substrate, which is enough to illuminate an
array of THz lenses. To detect the transmitted power, a receiver is placed just behind the lens in
order to minimize losses. Using this measurement setup, several of the lenses fabricated were
characterized. To verify the lens performance, we first measured the far-field setup without a
lens to collect reference data. The transmission characteristics of lens are recorded by placing the
lens array in between the transmitter and the receiver.

Figure 6.7 Block diagram of far-field measurement setup.
119

A second setup, near-field measurement, was used to characterize individual lens
elements. Figure 6.8 shows the layout of the near-field measurement setup. In order to measure
the position of high received power intensity, planar scanning method was adopted. To limit the
transmission, the metal plate (Cu, t=450 μm) with aperture, was placed in between a lens and a
receiver. The diameter of the aperture was designed to have a cut-off frequency below the design
frequency (f = 300 GHz). This metallic aperture structure plays a role as a near-field probe,
because it can scan and measure the power density on the surface of a lens. An absorbing
aperture material instead of Cu would be desirable to minimize multiple reflections. This will be
utilized in future experiments. A 2-axis robotic manipulator was used allowing a step resolution
of 200 μm on each axis. The receiver behind the metallic aperture measures the received power
density passing through the dielectric lens. The lens was fixed, whereas the aperture was attached
to the robotic arm. The receiver was also fixed at it has a wide capture window and not necessary
to move. Through this setup, spatial power density across the lens array was measured. For the
first measurements, on scanning in one axis was carried out with a resolution of 1 mm between
the data points.

120

(a)

(b)
Figure 6.8 Block diagram and a photo of planar scanning for near-field measurement setup.

121

6. 5. Measurement results
In order to characterize the electrical properties of a dielectric material that was used in
the fabrication lenses at THz frequencies; this paper adopts and utilizes the material
characterization method as described in [43]. The measured properties for a candidate material
are shown in Figure 6.9. The dielectric constant of the material ranges from 2.73 to 2.81 in THz
frequency range (up to 0.8 THz). Loss tangent (tan δ) of the material increases linearly from
0.012 to 0.05 from 0.1 THz to 0.8 THz, respectively.
The measured result of near-field planar scanning is shown in Figure 6.10. The step
size of axis-movement is set to a millimeter unit (d=1 mm) and the total scan length is 11 mm
with a metallic aperture (Adia=1 mm). The data shown in Figure 6.11 is based on the measured
results of near-field scanning of an extended hemispherical lens (D=5 × λ0). The two strongest
hot spots were examined in measured results. The result shows that the distance between two hot
spots is equal to the distance between the unit lenses, 5mm. This suggests that the hot spot of the
individual lens, as simulated above, are arrayed similarly as the lenses. The focal spot is smaller
than the aperture window and thus the measured result is spread of over a large window than
calculated through simulations. Actual intensity difference between the higher peak and valley
region will be higher if measured using a smaller aperture window.

122

2.82

Dielectric constant

2.80
2.78
2.76
2.74
2.72
0.1

0.2

0.3

0.4

0.5

0.6

0.7

0.8

0.6

0.7

0.8

Frequency (THz)

(a)

0.05

Loss tangent

0.04

0.03

0.02

0.01
0.1

0.2

0.3

0.4

0.5

Frequency (THz)

(b)
Figure 6.9 The (a) dielectric constant and (b) loss tangent of ABS material used in the fabrication
of lenses.

123

(a)

Measured power density

Magnitude (a.u.)

0.20

0.15

0.10

0.05
0

2

4

6

8

10

Distance (mm)
(b)
Figure 6.10 Measured results of Near-field planar scanning: (a) Schematics of scanning, (b)
Measured intensity plot across the lens array (across two lenses).

124

Figure 6.11 shows the measured transmission coefficient of an extended hemispherical
lens (D=5 × λ0) using the far-field measurement setup. In order to compute the normalized
transmission coefficient with a given lens, a reference coefficient is obtained first, defined by the
far field characteristics of the testing configuration without a lens. Then the far-field properties
of fabricated lenses were examined. In Figure 6.11 (a), it is clear the maximum transmission
coefficient is 0.48 and an overall transmission coefficient (> 0.3) is presented between 0.13 and
0.28 THz, suggesting a 0.15 THz usable bandwidth. Figure 6.11 (b) shows the response obtained
from another fabricated extended hemispherical lens (D=10 × λ0) showing that the measured
usable bandwidth for this lens is 0.1 to 0.18 THz. It should be noted that the measured center
frequencies are different from the desired center frequencies of both of designed THz lenses.
This is due to the receiver being placed on a slightly different position compared to the
calculated focal point.

125

1

Extended hemispherical(D=5 X 0)

Normalized transmission coefficient

0.9
0.8
0.7
0.6
0.5
0.4
0.3
0.2
0.1
0
0

0.1

0.2

0.3
Frequency [THz]

0.4

0.5

0.6

(a)
1
Extended hemisphercial (D=10 X  )
0

Normalized transmission coefficient

0.9
0.8
0.7
0.6
0.5
0.4
0.3
0.2
0.1
0
0

0.1

0.2

0.3
Frequency [THz]

0.4

0.5

0.6

(b)
Figure 6.11 Far-field measurement results of extended hemispherical lenses; (a) D=5 × λ0, (b)
D=10 × λ0.

126

Figure 6.12 shows the transmission responses obtained from an extended hemispherical
lens and a hypo-hemispherical lens with same diameter, D=5 × λ0. In this measurement, both of
the lenses show analogous transmission patterns. This is because the curvilinear contour of both
lenses was designed with the same diameter. Above 0.3 THz, the hypo-hemispherical lens
showed more high transmission coefficient. This is because a hypo lens is thinner than that of an
extended hemispherical lens. This degradation of transmission can be improved by using low
loss polymer materials such as HDPE and LDPE instead of ABS.

1

Extended hemispherical (D=5 X 0)

Normalized tranmission coefficient

Hypo-hemispherical (D=5 X 0)
0.8

0.6

0.4

0.2

0
0

0.1

0.2

0.3
Frequency [THz]

0.4

0.5

0.6

Figure 6.12 Measured transmission coefficients of two different lenses with the same diameter.

127

6. 6. Conclusions
In this work, THz dielectric lens array have been designed, fabricated and characterized
over a wide frequency range. Fabrication was carried out using a 3D plastic printer using ABS
dielectric material. This process is suitable for the fabrication of lens array. It can be utilized in
the direct fabrication of lens at the wafer level. Fabricated lenses have been characterized using a
simple near and far field measurement setup. Measured results clearly exhibit the expected
transmission characteristics of lenses. The proposed approach can reduce complexity in assembly
and is simple to implement.

128

CHAPTER 7. THZ IMAGING ARRAY SYSTEM
7. 1. Design and simulation
7.1.1 Microbolometer based on metamaterial structure
One of the key advantages of using metamaterials is that the frequency of operation can
be changed arbitrarily through manipulation of geometry of a unit cell. Here a modified slit ring
resonator (SRR), [77 - 79], is adopted in the design of unit cells operating at 0.1 and 0.4 THz.
These cells are scaled version of each other. Geometry of each unit cell is optimized to achieve a
compact sub-wavelength size structure. Figure 7.1 shows the design layout of the unit cells.
Figure 7.1 (a) is the basic model of the SRR, it is disjointed from its neighboring cells. Figure 7.1
(b) and (c) shows the unit cell which is composed of interconnected metamaterial unit cells. The
unit cells are connected together so that the resistance change can be directly measured across
these groups of cells. These unit cells are connected together using two approaches: straight and
zigzag. In the straight connection approach, the top sections of the unit cells are connected
together. In the zigzag approach, the diagonals of the unit cells are connected together. The
zigzag approach provides higher resistance path. Table 1 summarizes the physical dimensions of
the unit cell and array. The periodicity of the unit cells is approximately one sixth of the free
space wavelength at these design frequencies.

129

(a)

©
(b)
Figure 7.1 The geometry of SRR structure, (a) a modified SRR unitcell, (b) a straight
connection unitcell, and (c) a zigzag connection array of unitcells.

130

Table 7.1 Dimensions of a unitcell of designed microbolometers
Dimensions
Parameters

Remarks
0.1 THz

0.4 THz

Periodicity (P)

0.47 mm

0.115 mm

Width (W)

0.415 mm

0.105 mm

length of arm (Larm)

0.135 mm

0.035 mm

Width of arm (D)

40 μm

10 μm

Gap (G)

20 μm

5 μm

- λ0.1 THz= 3 mm (λ0.1 THz /P=6.38)
- λ0.4 THz=0.75 mm (λ0.4 THz /P=6.52)

Ansoft HFSS v.14, a full wave EM simulator based on finite element method (FEM),
was utilized in the design of unit cell structures. A low loss polymer substrate, PEEK (polyether
ether ketone), is used as a substrate material. The substrate thickness is 250 µm and its electrical
properties in the THz spectral region are presented in [43]. The dielectric properties of PEEK are:
dielectric constant, εr=3 and loss tangent, tan δ = 0.05 over a frequency range of 0.1 to 0.6 THz
[43]. PEEK is inert to most chemicals used in microfabrication, has low moisture absorption and
low thermal conductivity. A solid ground plane (Cu, 1 μm thick with thin Ti adhesive layer) is ebeam deposited on one side of the substrate. A 500 nm thick e-beam deposited nickel (Ni) is
used in the patterning of the absorbing structures. The thickness of this conductor is less than the
wavelength at 0.1 THz and it is a few times thicker than its skin depth.
Figures 7.2 (a) and (b) shows the simulation results of unit cells of microbolometer
operating 0.1 and 0.4 THz, respectively. The proposed SRR is dependent on the polarization of
incident waves. Simulations are carried out for both vertical (V-pol) and horizontal polarization
(H-pol) incident waves. For the analysis, the array is assumed to be infinite in periodicity and
thus Floquet’s theorem analysis is carried out on HFSS.
131

1.0

-5

0.8

-10

0.6

-15

0.4

-20

S11(V-pol)
S11(H-pol)
Absorption(V-pol)
Absorption(H-pol)

-25
50

100

150

200

Absorption

S11 (dB)

0

0.2

0.0
250

Frequency (GHz)

(a)
0

1.0

-2
0.8

0.6

-6
-8

0.4

Absorption

S11 (dB)

-4

-10
S11(V-pol)
S11(H-pol)
Absorption(V-pol)
Absorption(H-pol)

-12
-14
300

400

500

600

700

0.2

0.0
800

Frequency (GHz)

(b)
Figure 7.2 Simulation results (reflectivity and absorption) of a proposed unit cell of imaging
array, (a) V- and H-polarization of 0.1 THz unit cell and (b) V- and H-polarization of 0.4 THz
unitcell.

132

In case of vertical incident wave, the resonant frequency of the structure is excited at
0.1 THz. The reflected signal (S11) is approximately -18 dB and this translates to 98.5 %
absorption. The absorption is simply calculated by subtracting the reflected power from unitary
incident power. The power transmitted through the sample is negligible as there is a solid ground
plane present on the opposite side. The usable bandwidth (above 50 % of absorption) of the unit
cell (V-pol) is 26 GHz (90 to 116 GHz). In the case of H-pol incident wave, the resonant
frequency is excited at 0.183 THz. This is different from the excitement of V-pol resonance
frequency because of difference in electrical lengths (shown in Figure 7.3 of surface current
distributions).

V-pol (TE)

H-pol (TM)

(a)
Figure 7.3 Surface current distributions of each excited mode, (a) Magnitude and (b) Vector of
surface currents, V- and H-pol (from up and down).

133

Figure 7.3 (cont’d)

V-pol (TE)

H-pol (TM)

(b)
The minimum S11 was calculated to be -24 dB at resonant frequency of 0.183 THz. The
maximum calculated absorption is 99.5 % at this resonance frequency. The usable half power
absorption bandwidth was calculated to be approximately 54 GHz (between 0.16 to 0.214 THz).
Similar design and analysis was carried out for a scaled 0.4 THz unit cell structure. For
the physical parameters tabulated in Table 1, a 0.394 THz resonance frequency is calculated for

134

V-pol mode excitation. The reflected signal, S11, is determined to be -9.5 dB. This translates to
absorption of 88.7 % at the resonant frequency. The bandwidth of 0.4 THz unit cell (V-pol) is
127 GHz (between 0.350 to 0.47 THz). For H-pol mode excitation, the resonant frequency was
calculated to be 0.736 THz and the reflected signal (S11) is approximately -13.4 dB.

135

7.1.2 Beam focusing element- dielectric lens
The lens designs were carried out using the design approach outlined in Chapter 6. The
operating frequency is chosen to be 0.1 THz, same as the operating frequency of the
microbolometer. The simulated results of the lens are shown in Figure 7.4. The lens is composed
of the hemispherical part and the extended part. As mentioned in the previous chapter, this
configuration has an advantage that it can easily be integrated with microbolometers. Figure 7.4
(a) shows the layout of the extended hemispherical dielectric lens. The physical dimensions of
the extended lens are the radius (R) = 9.88 mm and the extension length (L) = 3.85 mm,
respectively. For the fabrication, plastic material (Acrylic, εr=2.7) was considered. This material
is not only suitable for three dimensional printing, but also shows the low loss at THz frequency
range. All physical values of the lens are designed to integrate this structure with the
microbolometer array. The lens diameter is six times larger than the wavelength of the operating
frequency. Figure 7.4 (b) shows the simulated far-field radiation of E-field. The radiation shows
the planar wave through the lens on the exciting wave. The waveport is referred to the dimension
of a W-band waveguide, the size of a port is 1.6 mm × 1.6 mm, because the wanted frequencies
travels well through the waveport and excites and couples to the lens. Also, as increases the port
size same as the initial designed array size (area of 3 × 3 array =5.4 mm × 5.4 mm), about 0.55
cm × 0.55 cm, the radiation wave shows the good pattern as expected. Figure 7.4 (c)
demonstrates the simulation result of the far-field gain of a dielectric lens. The calculated gain
was obtained as 15.5 dB. This performance is similar with lenses to analyze and characterize the
chapter 6. The radiation patterns of the lens are described in Figure 7.4 (d). Both radiation
patterns of E- and H-plane are well matched with the typical pattern of the dielectric lens with a
square wave radiating port.
136

(a)

(b)
XYPlot 3

NO_coating_discrete

ANSOFT

20.00
Curve Info
dB(GainTotal)
Setup1 : LastAdaptive
$coat='0.58mm' $e='3.85mm' $r='9.88mm' Freq='150GHz' Phi='0deg'
dB(GainTotal)
Setup1 : LastAdaptive
$coat='0.58mm' $e='3.85mm' $r='9.88mm' Freq='150GHz' Phi='90deg'

dB(GainTotal)

0.00

-20.00

-40.00
-90.00

-45.00

0.00
Theta [deg]

45.00

90.00

(c)
Figure 7.4 Schematics and simulation results of designed dielectric lens, (a) layout, (b) far-field
radiation (E-field), (c) gain, and (d) radiation pattern of the lens.
137

Figure 7.4 (cont’d)
Radiation Pattern 2
Curve Inf o
dB(GainTotal)
Setup1 : LastAdaptive
Freq='150GHz' Phi='0deg'

0
-30

10.00

30

dB(GainTotal)
Setup1 : LastAdaptive
Freq='150GHz' Phi='90deg'

0.00

-60

60

-10.00
-20.00
-30.00

-90

90
(d)

-120

120

-150

150
-180

138

7. 2. Imaging array layout design
Previous sections, the frequency dependent unitcell for detecting the incident power and
beam focusing element, the dielectric lens have been investigated. The goal of this dissertation is
to be introduced the multi-unitcell array for imaging system. Thus, the array design must
consider the arrangement of proposed microbolometer based on metamaterials working at THz
range. Figure 7.5 shows the unitcell alignment to construct the imaging array system. Each pixel
is composed of 4 by 4 unitcells. The design of a pixel helps the detecting capability improve
because the total resistance of pixel can increase to connect series. The geometry of a pixel is
also considered to measure the variation of resistance. One side connects to the signal pad
located on outer, and the other side goes to the ground pad shown in Figure 7.5 (a). Figure 7.5 (b)
and (c) demonstrate the total layout of proposed THz imaging array systems. The straight
connection array consists of nine microbolometers, nine signal pads, and two common ground
pads as shown in Figure 7.5 (b). The zigzag connected imaging array, shown in Figure 7.5 (c) is
the layout same as the straight connection array.
Pads shown in above figures are composed of copper having a good electrical conductor.
The square shape of a pad is for measuring the alteration of surface resistance each pixel. The
ground plane is produced in two circular pads on the designed layout. Total size of the 3 × 3 THz
imaging array is 2 cm × 2cm (including test pads), the detecting area where is placed pixels is
approximately 7.5 mm × 7.5 mm. Fabrication of this structure requires two mask layers. The first
mask layer is to pattern the Ni layer and the second mask is needed to pattern the Cu pads and
traces.

139

(a)

(b)

(c)
Figure 7.5 Layouts of 0.1 THz imaging array pixels, (a) configuration of a pixel, (b) the straight
connected imaging array, and (c) the zigzag connected imaging array.
140

7. 3. Fabrication
In order to fabricate designed THz imaging array, the microfabrication process was
used to implement the THz imaging pixel array. First step of the process, using acetone,
methanol, and isopropanol, the substrate that used same as the simulation considered, LCP and
PEEK should be cleaned to remove the residual particle and dust on. After drying in an oven,
photoresist, Shipley 1813 positive photoresist, is deposited on the substrate to use a spin-coater.
The speed of the spin-coater is set as 4000 rpm, and the time set as 30 sec. This can be achieved
with 1.5 μm of the photoresist thickness. After deposit the photoresist, the baking should be
conducted in the hot oven or on the hot plate for 10 minutes (hot oven) or 1 min (hot plate). Then,
using UV source, the lithography has been carried out for 1.1 min. In development process, to
immerse the illuminated sample in MF 321 developer for 25 sec and 10 sec in a row, also in
order to get the best results, the brief oxygen plasma “de-scum” was used between lithography
and metal deposition [80 - 82]. This de-scum process will remove any thin photoresist residue
that could cause poor metal adhesion or contact. A 120W, 30 second oxygen plasma at 100% O2
flow rate (all other gases off) should be sufficient. The e-beam evaporator was utilized to deposit
the first layer metal, nickel on top of the photoresist. Metallic layers are composed of double
layers using different metals. In order to enhance the adhesive metal deposition, titanium has
been used for an adhesive layer (t= 300 Å). The deposition ratio of nickel was set as 0.5 Å/sec to
metallize the nickel pattern to make even surface roughness and 5,000 Å of final thickness of
nickel was achieved. The metal liftoff was performed with a sonicator. The sonicator help
decrease the lift-off time required for the acetone soak. However, the sonicator doesn’t use over
5 min to avoid the metal stress due to thin (< 1 μm) metal film. Once the lift-off is completed,
washing it with isopropanol and agitate for 1 minutes and rinse with DI water and dry out with
141

N2 gas. To produce second layer of the THz imaging array, follow the same procedures such as
spinning coating, lithography, and developing. On photoresist deposition, to adjust the spinning
speed 4,000 to 2,000 rpm, this setting helps the thickness of photoresist deposit up to 2 μm. This
thickness supports to help the second metal lift-off process. The sputtering has been used to
deposit the second metal layer (copper) on top of the first layer. Also, using the lift-off process
mentioned above, the unwanted pattern metal layer is removed. The detail process is described in
Figure 7.6.

Figure 7.6 Microfabrication process of a proposed THz imaging array.

142

Figure 7.7 shows example photographs of the fabricated THz imaging array. The
physical dimension of the fabricated nickel microbolometer pixel array was inspected using a
microscope. As shown in pictures, the metal patterns and interconnections were well aligned the
exact positions that designed on mask layouts.

Figure 7.7 Microphotographs of manufactured THz imaging array.

143

7. 4. Experimental measurement setups
Two measurement setups are used in the characterization of the pixel elements and the
imaging array. First setup is made to determine the absorbed power by the pixel elements and the
second setup is made to characterize imaging characteristics of the fabricated devices.
The setup for the measurement of power absorbed depicted in Figure 7.8 (a). In this
setup, an array of absorbing elements is placed in front of the horn antenna. A W-band
directional coupler is attached between the horn antenna and the backward wave oscillator
(BWO, W-band source). A W-band receiver (a calibrated harmonic mixer) attached to the
directional coupler was used to measure the reflected power. For calibration, a flat metal sheet
was used that was placed at the same distance as the absorbing elements. The reflected signal
from the reference and the array elements was measured and the difference in signal is used to
calculate the absorbed power.
An open-ended W-band waveguide was used to illuminate the object to be imaged.
Measurement was setup for capturing the image in the transmission mode depicted in Figure 7.8
(b). Since the array is composed of 3×3 array, a 2-axis robotic manipulator was used with step
resolution of 200 μm on each axis to raster the sample to be able to capture a complete image of
the sample. A multimeter along with data acquisition was used to measure the resistance across
the individual pixel elements. The THz imaging array was fixed, whereas the object under scan
was attached to the robotic arm. Through this setup, change of surface resistance of each pixel
across the array was measured to capture the image.

144

(a)

Figure 7.8 Measurement setup block diagrams, (a) Absorption measurement setup, and (b)
Imaging measurement setup.

145

Generally, the noise equivalent power (NEP) of a bolometer is defined as the power
needed to achieve a signal to noise ratio of unit at 1 Hz bandwidth [83]. To indirectly measure
the intrinsic NEP of the bolometer, a bias circuit with a DC voltage and a load resistor was used.
In this configuration [28, 29], a voltage is applied across the series connected bolometer and
fixed temperature insensitive resistor. A voltage drop is measured across the bolometer as a
function of applied current. The detail layout of a measured setup is shown in Figure 7.9.

Figure 7.9 Measuring setup of noise equivalent power.

146

7. 5. Measurement results
Absorption measurements were carried out from 85 to 105 GHz with step resolution of
1 GHz. Figure 7.10 shows the measured result of the proposed THz microbolometer array. In
order to characterize the absorption of the bolometer array, the main orientation of polarization,
vertical polarization was excited. The measured results matches closely with the simulation
results (V-pol: 0.1 THz at Figure 7.2 (a)). The results show strong absorption for V-pol from 87
GHz to 107 GHz (shown in Figure 7.10). The absorption is determined to be 9.8 dB at 102 GHz.
It is lower than the simulated absorption of 18 dB. Rest of frequency range could not be
measured as the source frequency is limited to 110 GHz. However, we can estimate that the
performance of rest of frequencies will have similar results as the simulation results shown in
Figure 7.2 (a).

0

1.00
dB
ratio

2

0.95

6
8

0.90

10
12

0.85
85

90

95

100

105

Frequency (GHz)

Figure 7.10 Measured result of power absorption of THz imaging array.

147

Absorption

Absorption (dB)

4

Figure 7.11 shows the measured resistance of a bolometer versus applied current while
the environment temperature is fixed (T=296 ⁰K). The power drop across the bolometer converts
to heat and which in turn increases the resistance of the bolometer as seen in Figure 7.11 (a).
This can be plotted as a function resistance versus power drop across a bolometer as shown in
Figure 7.11 (b). The slope of Figure 7.11 (b) gives the sensitivity of the bolometer. NEP of the
bolometer can be calculated using the slope (sensitivity), TCR of the film and Johnson noise [29,
83, 84]. The TCR of Ni was introduced in Figure 4.6 with similar thickness processed using e-3

beam (α=2.7×10

-1

grad ). Using these measure results and supposing the Johnson noise is the

only parameter defining the radiation equivalent to the power of the NEP, the calculated intrinsic
-9

NEP of a bolometer is 1.652×10

W/Hz

1/2

. Considering to the absorption property of the
-9

bolometer, the estimated the NEP of a sample can be analyzed 8.26×10

W/Hz

1/2

at 100 GHz

due to the frequency shifting. The detailed parameters that used to calculate the NEP are shown
in Table 7.2.
Table 7.2. NEP calculated parameters
Parameters

value

Remarks

R0=49.153 Ω

No bias

R1=166.3 Ω

I=17 μA
V=13V

Resistance (R)

Power (W)

82.5 μW

Responsivity (S)

0.997 V/W

148

200
180

Resistance ()

160
140
120
100
80
60
40
0.0

0.1

0.2

0.3

0.4

0.5

0.6

0.7

0.8

Current (A)

(a)

200
180
Resistnace ()

160
140
120
100
80
60
40
0

20

40

60

80

100

Power (W)

(b)
Figure 7.11 Measured resistance versus (a) current and (b) power.
149

120

In order to demonstrate the viability of the proposed imaging array, an imaging scan
was carried using the fabricated structures. Figure 7.12 (a) shows a photograph of a sample with
punched holes in a metallic sheet. The minimum size of holes is 0.9 cm × 0.9 cm which is
approximately twice the size of a pixel element on each side. Figure 7.12 (b) shows the
transmitted signal as a function of position across the sample. The measurement step of X- and
Y-axis robotic arm set at 6mm in each direction. A total of 11 × 11 = 121 points are measured
over the XY plane and the spacing between the source and the detector was set to approximately
7cm.
The setup shown in Figure 7.12 (b) can be used in transmission measurements and
shapes of structures can be resolved using a bolometer pixel element. Scattering at the edge of
punched holes makes the holes appear larger than their actual size. For the measurement of an
actual sample hidden in a paper envelope the setup shown in Figure 7.13 (a) was used. Here, to
minimize scattering and to attain a close to a plane wave impinging on a sample an open ended
WR-10 waveguide at the source end was used in place of the horn antenna. Furthermore, the
raster steps were decreased to improve resolution. Measured image of a metalized sample placed
in an optically opaque envelope between the source and the imaging array is shown in Figure
7.13. In this picture, colors represent the density of the signal after passing through the sample.
Blue color represents low-transmission and red color represents regions with high transmitted
signal. The physical size of the sample is approximately 2.8 cm × 2.5 cm. In this measurement,
the axial step size is set as 2mm per step. A total of 21 × 21 = 441 points are collected in the XY
plane. The image scanned clearly shows the hidden object. This clearly shows that proposed
imaging array of this paper can be used in high sensitivity imaging applications.

150

(a)

(b)
Figure 7.12 Measurement setup (a) and scanned result (b) of punched holes on the metallic sheet.

151

(a)

(b)
Figure 7.13 Scanning layout and results of concealed S-shaped metal, (a) scanning layout, (b)
scanned metallic S-shape pendant, (c) and (d) scanned results with different scanning
resolution.(set raster scanned resolution: (c) 4mm/step and (d) 2mm/step).

152

Figure 7.13 (cont’d)

10
0.1720
0.1970
0.2220
0.2470
0.2720
0.2970
0.3220
0.3470

Y-axis (4mm/step)

8

6

4

2

0
0

2

4

6

X-axis (4mm/step)
(c)

(d)
153

8

10

7. 6. Further improving NEP of the bolometer
We discussed about the new THz imaging array in previous sections. The most
important parameter, the NEP of the bolometer which is mainly composed of the imager, must
be considered to develop the imaging array. In this section, we would introduce the way to
improve the NEP of the bolometer further. Normally, the thermal bolometer has a noise that is
only related to the variation of the heat loss to the heat sink, and the thermal coupling radiates
purely [28, 29, 83]. Thus, the NEP is derived by

√

7 – (1)

2

-8

Where A is the area of a pixel (m ), σ is Stefan-Boltzmann constant (=5.67×10
2

4

-23

J/(m ∙S∙K ), k is Boltzmann constant (=1.374×10

J/K), T is the temperature (°K), and ϵ is the

emissivity [83]. From this equation, the NEP could be manipulated by adjusting parameters of
the equation. For example, as decreasing the effective area of a pixel or increasing the
absorptance of a bolometer, the value of NEP goes smaller, it can be achieved the higher
sensitivity of the bolometer.
Our approach originates from miniaturizing the effective area of a pixel. As similar as
the proposed metamaterial bolometer, other candidating models based on modified slit ring
resonator (SRR) structure are considered to build up the low NEP compact bolometer [85].
7.6.1 NEP analysis
In this section, the intrinsic NEP of the bolometer will be introduced. From the equation,
7 – (1), we can estimate the intrinsic NEP of the ideal bolometer. This computed NEP is used as
a reference value to compare the performance of practical bolometers. It has been indicated that
154

this reference to assume a perfectly noiseless environment of the bolometer [86]. Here is an
example to analyze the NEP of a bolometer. Assume a blank metal layer that is composed of an
absorbing layer on a typical infrared bolometer. The amount of absorbing incident wave by the
bolometer is considered as 0.5 (equal to 3dB of the reflectivity). Figure 7.14 (a) shows the
calculated intrinsic NEP according to different physical size of a bolometer. In this computation,
we assume that the bolometer is operated at room temperature and the physical size is a 50 μm ×
50 μm of a bolometer. The intrinsic NEP of it would be calculated as 3.16 × 10

-13

W/Hz

1/2

at

room temperature. Under same condition, when the dimension has been changed, the NEP is
analyzed to increase the NEP. It means the performance of the bolometer to degrade. Another
example, there are two bolometers to have same physical dimension (300 μm × 300 μm), only
difference between two samples is the absorption (ϵ), one is 0.5 and the other is 0.95. In this case,
-12

the higher absorption (ϵ = 0.95) bolometer shows the lower NEP (=1.37 × 10

W/Hz

1/2

,

depicted red dot in Figure 7.14 (a)) compared to the lower absorption (ϵ = 0.5) bolometer (=1.9 ×
-12

10

W/Hz

1/2

), approximately 72 % degradation. Through, this analysis, we can say that higher

absorption bolometer is able to work better performance than the lower and verify our approach,
high absorption and small effective area to enhance the bolometer performance by this analysis.
Figure 7.14(b) shows the relationship between temperature and the NEP. Compared to
the variation of NEP dependent on dimension, it of NEP dependent on temperature changes
rapidly, roughly seven order of the calculated value. It is because the main factor of the equation,
7-(1) is temperature (T) having the variation of temperature to the fifth power.

155

NEP (W/Hz1/2)

-12

10

-13

10

0

200

400

600

800

1000

Dimension (m)

(a)

-12

10

NEP

-13

10
NEP (W/Hz1/2)

-14

10

-15

10

-16

10

-17

10

-18

10

-19

10

0

100

200

300

400

500

600

700

800

Temperature (K)

(b)
Figure 7.14 Analysis result of NEP, (a) dimension dependent and (b) temperature dependent.
156

7.6.2 Miniaturized metamaterial (U-shape) bolometer
The physical dimensions of a miniaturized bolometer are the periodicity (P) of a unitcell
is 360 μm, the length (L) of a unitcell is 298.6 μm shown in Figure 7.15 (a) [85, 87 - 90]. In
order to improve the absorption of the proposed bolometer, we investigate the relation between
the number of the arm and the amount of absorption. The width (w) and gap (g) of a unitcell are
designed as 20 μm and 18 μm, respectively (depicted in Figure 7.15 (a)). The spacing (S)
between arms is set as 10μm. Schematics of absorbers that are adjusted the number of arms are
described in Figure 7.15 (b), (c), and (d). The detail physical information of the unitcell is
summarized in Table 7.3. The entire geometry of the proposed bolometer is composed of a
patterned conductor layer (top layer), dielectric substrate (middle layer), and ground plane in side
of a substrate (bottom layer). PEEK is adopted as a substrate same as introduced sample
discussed in previous sections. The conductor material is considered as nickel (Ni) to simulate
and fabricate the proposed absorber.

Table 7.3 Physical dimensions of the miniaturized (U-shape) bolometer
Physical dimensions

values

Remarks

P

360 μm

L

298.6 μm

g

18 μm

- Metal: nickel (Ni, 500 nm)

W

20 μm

- λ/P = 8.33 @ λ =3mm

S

20 μm

- Frequency: 100 GHz
- Material: PEEK

157

(a)

(b)
(c)
(d)
Figure 7.15 Schematics of the miniaturizing (U-shape) bolometer structure, (a) single arm, (b)
double arms, (c) triple arms, and (d) quad arms.

The proposed bolometer is analyzed EM properties by using HFSS as the same manner
as absorber that studied in previous chapters. Figure 7.16 and 7.17 demonstrate simulated
absorption properties of a U-shaped bolometer with vertical and horizontal E-polarization,
respectively. In Figure 7.16, the reflectivity of a single arm is computed as 3.15 dB at 103.1 GHz
that is equivalent to the converted absorption, approximately 0.52. In case of a U-shape
bolometer having quad arms, the reflectivity shows -14 dB at 109.7 GHz (absorption = 0.96).
Simulated results of the bolometer having double and triple arms describe high absorption (0.92
and 0.95).

158

0

Reflectivity (dB)

-5

-10
single
double
triple
quad

-15
50

100

150

200

Frequency (GHz)

(a)
1.0

Absorption

0.8

0.6

0.4
single
double
triple
quad

0.2

0.0
50

100

150

200

Frequency (GHz)

(b)
Figure 7.16 Simulated results of a miniaturized (U-shape) absorber (V-pol.), (a) reflectivity and
(b) absorption.
159

0

Reflectivity (dB)

-5

-10

-15
single
double
triple
quad

-20

-25
50

100

150

200

Frequency (GHz)

(a)
1.0

Absorption

0.8

0.6

0.4
single
double
triple
quad

0.2

0.0
50

100

150

200

Frequency (GHz)

(b)
Figure 7.17 Simulated results of a modified absorber (H-pol.), (a) reflectivity and (b) absorption
ratio.

160

Usable absorbing half power bandwidth of each absorber with designed arms (single to
quad) are computed as 10.1 GHz (98.5 ~ 108.6 GHz), 50.1 GHz (87.2 ~ 137.3 GHz), 86.1 GHz
(85.4 ~ 171.5 GHz), and 92.4 GHz (84 ~ 176.4 GHz), respectively. According to this analysis,
we could conclude the relationship between numbers of arms, to increase the number of arm, to
enhance the absorption at vertical E-polarization. It is because the enlarging effective resistant
area helps produce higher surface resistance with increasing the number of arms. For horizontal
E- polarization, Figure 7.18 describes the computed results for reflectivity and absorption.
Compared to V-pol, the results demonstrate that increasing the number of arms makes the
operating frequency shift to higher frequency and reduces absorption.

161

7.6.3 Miniaturized metamaterial (hybrid serpentine/spiral) bolometer
In this section, we introduce a different type of metamaterial bolometer by modifying
the U-shaped structure discussed in the previous section. The schematic of this bolometer
resembles a hybrid of serpentine and spiral inductor structure. Dimensional information of a
miniaturized unitcell is the periodicity (P) of a unitcell is 194 μm, the length (L) of a unitcell is
180 μm shown in Figure 7.18 (a). The width (w) and gap (g) of a unitcell are designed as 12 μm
and 11 μm shown in Figure 7.18 (a). The spacing (S) is set at 11μm. Figure 7.18 (b) and (c)
shows the simulated results of a miniaturized structure that is designed to operate at 0.1 THz. It
shows strong absorption peak at approximately 98.4 GHz (in case of H-polarization). At this
frequency, the return loss (magnitude of S11) is approximately 12.4 dB. This translates to an
absolute absorption density of 94.3 %. From the simulations, the half power bandwidth (HPBW)
of the absorbing peak range from 71.4 GHz to 200 GHz. This structure provides a wide band
absorption which is desirable for many applications.

(a)
Figure 7.18 Simulated results of a proposed serpentine absorber, (a) schematic, (b) reflectivity,
and (c) absorption ratio.
162

Figure 7.18 (cont’d)

0
-2

Reflectivity (dB)

-4
-6
-8
-10
-12

H-pol
V-pol

-14
50

100

150

200

Frequency (GHz)

(b)

1.0
H-pol
V-pol

Absorption

0.8

0.6

0.4

0.2

0.0
50

100

150

Frequency (GHz)

(c)

163

200

7.6.4 Fabrication
In order to fabricate the designed miniaturized absorbers, fabrication processes
discussed in section 7.5 is utilized here again. Figure 7.19 (a) shows the mask layout of
fabricated bolometers having different numbers of unit cells per pixel element, 1 unit cell to 4 ×
4 unit cells. Additionally, there are four different designs, based on U-shaped unitcell and hybrid
serpentine/spiral unitcell mentioned in the previous section. Fabricated bolometers and its
assembly is shown in Figure 7.19 (b). This configuration allows ease of soldering of wires
between bolometers and a testing die. This approach protects the bolometer and holds it flat. A
test –die produced on FR-4 substrate is used to complete the assembly [91].

(a)

(b)
Figure 7.19 Mask layout (a) and photographs of fabricated bolometers (b).

164

The lens array has been designed to enhance the density of incident power and also to
characterize the performance of the bolometer designs. Detail physical dimensions are the
diameter which is 6 mm, and the extension length is 0.7 mm. In order to achieve good alignment
between a bolometer and lens and connected wires, the leg (spacer) of a lens array has been
drawn to a height of 1 mm. This length is optimum for focusing as confirmed through the EM
simulation and allows assembly of the system. The effective receiving area, where is 1mm away
2

from a lens, is calculated as 1 × 1 mm . Designed lens array has been fabricated using 3-D
plastic printing (ABS plastic). Figure 7.20 demonstrates the layout and photographs of a lens
array and EM simulation results of the lens.

(a)

(b)
Figure 7.20 Layout and photographs of a lens array (a) and E-field simulation (b).
165

7.6.5 Measurement
In order to measure the bolometer and lens array, the near-field radiation setup is used.
The setup for near-field radiation measurement setup is shown in Figure 7.21 (a). Through this
measurement, the sensitivity of a bolometer has been characterized. Moreover, the sensitivity of
the integrated system, a bolometer and a lens array has been measured. First, the sensitivity of a
bolometer is tested using the configuration without a lens array. In this setup, each bolometer is
measured for variation in resistance under illumination using a multimeter. It is measured again
by the lens array, the lens array sits on top of a bolometer. Resistance change was measured
several times and the average values are presented.

(a)

(b)
Figure 7.21 Measurement setup for the assembly of a bolometer and a lens array.

166

Figure 7.22 demonstrates the measured result obtained using setup shown in Figure 7.21.
As shown in Figure 7.22, the graph plots the number of unitcell array versus the variation of
resistance. Also, it shows the effect of a polarization of incident wave and its effect on resistance
change upon illumination. As the number of unitcell increases per pixel the change in resistance
gets bigger for V-polarization illumination. The change in resistance increases when a lens is
introduced on top of the bolometer structure. The lens focuses larger amount of power onto the
bolometer structures.

Normalized variation of resistance

1.0007
V-pol (no lens)
H-pol (no lens)
V-pol (with lens)
H-pol (with lens)

1.0006
1.0005
1.0004
1.0003
1.0002
1.0001
1 unitcell

2 X 2 unitcells

3 X 3 unitcells

Bolometer

Figure 7.22 Measured results of resistance variation.

167

4 X 4 unitcells

CHAPTER 8. CONCLUSIONS AND FUTURE WORK
8. 1. Conclusions
In this dissertation, a novel THz imaging array has been proposed and presented. The
design of a bolometer is based on the use of frequency dependent periodic structures (FSS and
metamaterials) and high surface resistivity adopting the skin effect. Combining these properties,
high absorbing bolometers have been developed to replace the scheme of conventional imaging
arrays. In order to improve the density of incident wave, micro-optics has been utilized to be
suitable to integrate with planar bolometers. Plasmonic bulls-eye antenna and lens antenna have
been investigated to use new cost effective fabrication processes of the use of a wet etching and
3D printing manufacturing method. Based on these research achievements, the novel THz
imaging array system has been introduced and discussed about the integration way and the
characterization of the imager.
Various groups of thin-film THz absorbers using metamaterials have been investigated
through EM analysis, design and fabrication in chapter 3. Designs of THz absorber show strong
absorption and ultra-wide bandwidth at terahertz frequency ranges. The proposed approach takes
advantage of skin-effect losses in order to make high resistivity and wide band metamaterial
absorbers. To enhance sensitivity high resistivity metal (nickel) is utilized in the fabrication of
the absorbing structure. Further enhances in operating bandwidth were exhibited through multistacking of metamaterial absorbers. The results of proposed metamaterial absorbers clearly
describe a new way to achieve ultra-wide band absorption with strong absorption. THz
Absorbers having large usable bandwidth can easily be designed using the proposed approach.

168

The absorbers have been measured to use frequency and time domain measurement setup. These
absorbers can be applied in sensing and imaging systems.
Based on this research achievement, the new type of a power meter has been introduced
in chapter 4. Proposed periodic structures is to use to build up the power detector and sub
components such as interconnection between periodic structure and pads to connect with test
apparatuses to measure the characteristics of a power meter have been designed. Proposed power
meter designs have been analyzed to perform the EM simulation using HFSS. In order to
evaluate the detecting power sensitivity of a power meter, measurements are carried out to use
W-band high power BWO, largely restricted by existing measurement setup. Temperature
coefficient of resistance (TCR) of the nickel film is measured and shows high sensitivity to
change in temperature.
One of the advantages of decreasing the effective area of the bolometer is decrease in
noise and increase in sensitivity. However, with decrease in area the effective capture area
decreases. To overcome this challenge, micro-optics are used to focus the incoming radiation
onto the bolometer. Two types of micro-optics focusing elements are studied: i) dielectric lens
antenna and ii) bulls eye surface plasmonic antenna. The use of bulls-eye antenna element for
imaging array is demonstrated here for the first time. Details of the micro-optics are presented in
Chapters 5 and 6. Simulations of beam focusing elements were carried out using Ansoft HFSS.
For the fabrication of bulls eye structure was carried out using two approaches: i) wet-etching, ii)
3D plastic molding (printing). The dielectric lens antenna was fabricated using 3D plastic
printing. Integration of micro-optics at the wafer level is demonstrated using these fabrication
approaches. Measurements of the micro-optics are carried out in both far- and near-field regions.

169

In chapter 7, a new THz imaging array is proposed and demonstrated. The proposed
imaging array is based on high resistivity metamaterial that provides high absorption of incident
signal. The imaging pixel element is composed of an array of sub-wavelength metamaterial unit
cells. The unit cells can be scaled to design detector element in the desired band of operation.
Furthermore, the size of the unit cells can be reduced down into subwavelength size. These
subwavelength structures directly absorb THz radiation and this can be utilized as a detector
through a change in the resistance across the pixel element. An intrinsic NEP of 1.6×10
1/2

W/Hz

-9

is demonstrated at 0.1 THz. This can further be improved by optimizing thermal

isolation, TCR of the film and effective resistance of the film.
The entire results of this dissertation clearly demonstrate and verify that a novel THz
imaging array system, which is composed of new types of high resistivity frequency dependent
metamaterial absorbers, three dimensional high concentrating antenna elements (bulls eye and
lens) with various fabrication materials, and system level integrations, can be utilized to design
cost effective and simple THz imager applications.

170

8. 2. Future work
Future work could include further explorations in the development of miniaturizing the
unitcell, reducing the thermal loading between the absorber and substrate, utilizing high TCR
materials, and integration of the read-out circuitry. In this section, we would suggest the ways to
implement and improve the performance of a THz imaging array.
8.2.1 Genetic Algorithm
Recently, the use of genetic algorithms (GA) has become quite popular to optimize and
miniaturize antennas and RF/microwave circuits designs because GA well matched with existing
EM analysis techniques, and generally produces results that satisfy the given requirements [9294]. Many researches have already been carried out furthering both the computational maturity
of GA optimization in electromagnetics, as well as in expanding the domain of applications
including quite sophisticated designs [93], [95]–[96]. These research with GA optimization have
led many distinct results are novel configuration of cells and broadband/multiband operation.
Thus, in order to miniaturize the unitell, GAs can be utilized in future work.
Here is an example of GA optimization, filling beehive- cell (FBC) method, one of the
advanced structural design methodologies, originated from filling square-pixel (FSP) [94]. This
FBC can improve to organize several latticed shapes such as diagonal and rhomboidal. In this
analysis, using this GA method, the configuration of the cell can be manipulated as a variety of
filling shapes. Figure 8.2 demonstrates the samples of cell configurations based on FBC method
[94]. In order to generate the plotting function, FBC runs several iterations to optimize the
requirement using GA steps such as reproduction utilizing the biological operators came from
the theory of evolution, selection, survivor of the fittest position.

171

(a)

(b)

(c)

(d)

Figure 8.1 Various schematic of unicells generated from GA [94].
8.2.2 Thermal loading
Conventionally, for example, the infrared and mmW uncooled bolometers comprise of
thermally isolated layers deposited temperature sensitive materials on top [30]. Mostly, materials
can be utilized VOX, α-Si or poly-SiGe [83]. These thermal detecting layers are implemented to
use the microfabrication process directly on CMOS circuits. In this integrating configuration, for
optimum performance, the detectors must be thermally isolated from their auxiliaries [97].
Otherwise, the detecting element would become thermally loaded that it would harmfully affect
to degrade the function of power detection as thermal detectors.
Likewise, our bolometer consists of thin-film absorbers and a dielectric substrate.
Absorbers directly deposited on top of the substrate by using e-beam deposition. In this
configuration, despite of using low thermal conducting material, PEEK, there is large interfacing
area between absorbers and a substrate to worsen the sensitivity of a bolometer. Here are
alternative ways to improve the thermal loading. Firstly, the suspended scheme, well known as
RF-MEMS cavity, can be used instead of a substrate. To fabricate the cavity and a thin
membrane layer such as silicon nitride (Si3N4), the bolometer can be produced on top of a
silicon nitride layer and it help isolate thermal loading from surroundings. The other way is to
use a special material, Aerogel has the low thermal conductivity. Normally, Aerogel works as a
172

good thermal insulator because of their own characteristics to invalidate heat transfer methods
such as convection, conduction, and radiation [98]. Thus, the cavity can be filled with aerogel to
build up the thermal isolation layer. Figure 8.2 shows conceptual pictures of thermal isolation for
a bolometer. Figure 8.2 (a) demonstrate the scheme a suspended bolometer on the membrane and
a cavity fulfilled with air. The integrated pixel demonstrates in Figure 8.2 (b). A pixel consists of
2×2 unitcells on top of an aerogel cavity or silicon nitride membrane, and a readout circuit. Also,
there is a wire bonding to interconnect a bolometer and a readout circuits, which is needed to
measure the resistance variation of the detector element, which preclude from thermal leakage of
the absorbed radiation.

(a)
(b)
Figure 8.2 Conceptual pictures of thermal isolation, (a) cavity, and (b) integration a bolometer
with a readout circuit.

8.2.3 Readout circuits
Normally, the readout circuits are designed to assist a good interface between detecting
elements and the digital domain part such as signal processing [99]. In our case, we can adopt
two approaches here, Firstly, using commercial switching matrix (MUX), the readout circuit can
173

be built up easily. Figure 8.3 shows the proposed readout circuit to be able to measure a 3×3 THz
imaging array. It consists of a preamp to amplify the signal from a pixel, time division based
switching matrix, and data acquisition component. Using measured raw data, a scanned image
appears on display.

(a) Read-out circuitry
Figure 8.3 Proposed system level assembling methods and readout circuit.
Another approach is to use a variety of CMOS readout techniques such as sourcefollower per detector (SFD), direct injection (DI), and gate-modulation input (GMI) [99]. These
readout circuits can assist to develop the imaging system having large number of pixel, for
instance 100 × 100 array.
8.2.4 Temperature Coefficient of Resistivity (TCR)
We could use high TCR material to improve the sensitivity of the bolometer. Some
materials, for examples, nickel alloys, amorphous Si, and antimony (Sb) have higher TCR than
nickel. With this property, the bolometer can work more sensitive at the same amount of incident
power. It could assist lower NEP bolometer. Table 8.1 summarizes the TCR of various materials
[100,101].
174

Table 8.1 Temperature coefficient of resistance (TCR) of materials
-1

Material

Temperature Coefficient of Resistance (TCR, %K )

Copper

0.003862

Gold

0.0034

Nickel

0.006

Vanadium oxide (VOx)

0.021

Poly SiGe

0.02

α-Si:B:H

0.028

α-Si:H

0.039

GeSiOx

0.042

α-SiGe:HF

0.049

8.2.5 THz spectroscopy
Recently, spectroscopy is one of the most interesting applications of terahertz regimes
which grow from the strength of the signal source or absorption specification for a specimen
such as molecules [6].
We introduced many types of absorbing structure in this dissertation. From this, the
sensor for THz spectroscopy can be developed to modify and combine absorbing structure
together. Figure 8.4 (a) demonstrates the idea of the THz spectroscopy. In this combination of a
unitcell, there is two different size of double slit ring resonators, has yielded the multiple high
absorbing peaks. In order to verify this idea, the EM simulation has been carried out to use
Ansoft HFSS. Figure 8.4 (b) and (c) shows the simulation layout and the simulation results.
Especially, in Figure 8.4 (c), the multiple absorption peaks (above 70% absorption) are observed
175

at 0.1, 0.16, and 0.215 THz, respectively. In spectroscopy application, if the specimen is loaded
to the proposed sensor, the magnitude and frequency of absorption should change depend on the
material properties of the specimen.

(a)

(b)

0.8

Absorption

0.6

0.4

0.2
V-pol
H-pol

0.0
50

100

150

200

250

Frequency (GHz)

(c)
Figure 8.4 Simulation result for a muti-band unitcell, (a) layout of a unicell and (b) the
preliminary simulation result of absorption.
176

APPENDICES

177

APPENDIX A
Operative Microfabrication Procedures
A.1 Deep copper wet etching process
This etching process is a recipe of copper wet etching that developed by Terahertz
System Lab (Tesla) at Michigan State University. Using this proposed etching process,
microwave circuits, metallic bulls eye antennas and metallic molds of microfluidic capillaries
having the height of several tens to hundreds of micrometers have been fabricated.
Etching procedure steps are as follows:
1. Prepare the metallic plate or the substrate.
2. In order to remove contaminations on the surface of the substrate, standard RCA
cleaning or simple surface cleaning process (details to be introduced) are carried out.
3. Place the substrate in basket.
4. Spray and submerge the basket with the substrate in a beaker containing acetone for 20
seconds.
5. Remove the basket and the substrate from the acetone beaker and spray and immerse
them in a beaker containing Methanol for 20 seconds.
6. Remove the basket and the substrate from the acetone beaker and spray and immerse
them in a beaker containing deionized water (DI) for 1 min.
7. After cleaning the substrate, blow dry substrate with nitrogen gas (N2). The substrate
should be no stains on the surface.
8. Put the substrate in the hot oven at 95 °C for 5 min or bake the substrate at 105 °C for 1
min on the hot plate (this will assist to drive off remaining surface moisture to help
enhance the adhesion of photoresist (PR) to the substrate).
178

9. Using a plastic dropper, the substrate is coated with positive PR (Shipley 1813 in ERC
cleanroom).
10. Spin the PR coated substrate for 30 seconds at 4,000 rpm (t=1.2 μm).
11. Bake the substrate at 95 °C for 1 min on the hot plate or put the substrate in the hot
oven at 95 °C for 5 min.
12. Inspect the quality of coated PR. If there are some dusts or particle on the surface of PR,
go back to step 3 and redo the steps.
13. Use a mask aligner or photolithography machine to expose the PR under the set
parameters (275W, 1.1 min on Karl Suss MJB 3 mask aligner in ERC).
14. Submerge the substrate in the developer (MF-319) agitating continuously for 40 ~ 45
seconds.
15. Spray and immerse the substrate in a DI beaker in 1min.
16. Blow dry the substrate with N2.
17. Mix the wet etchant, DI: Sodium persulfate (Na2O8S2, MG Chemicals) =3:1(normally
1,000 ml: 330 g).
18. Heat up the wet etchant on the hot plate at 80 °C and measure the temperature of the
etchant up to 45 °C. After reaching the etchant temperature up to 45 °C, and then adjust
the temperature of the hot plate to maintain the etchant temperature 45 °C.
19. Immerse the exposed substrate into the etchant slowly.
20. The etch rate for copper by the mixed etchant is approximately 3.2 μm/min (Detail
measured etching rate shown in Figure A.1). After etching for the wanted amount of
time, remove the substrate from the wet etchant.
21. Rinse the substrate in DI water beaker for 1 min and blow dry it with N2.

179

22. Inspect the etched substrate under the microscope to investigate the etching pattern. If
the pattern has not been fully etched, go back to step 19.
23. Spray acetone on the substrate to remove the PR.
24. Rinse the final sample with methanol and DI water.
25. Finally, blow dry with N2.

Etching rate (um/min)

4

3

2

Etching rate

1

0
25

30

35

40

Temperature (C)
Figure A.1 The etching rate of the copper wet etchant.

180

45

50

A.2 De-scum process
This process will help prepare wafers and substrates for photolithography process may
need to remove photoresist residue in normally patterning (trenched) areas [102]. Usually, descum process is limited to the removal of photoresist by an oxygen or fluorine plasma, the
products being vapors or reactive species with photoresist to form ash which are volatile and
pumped out by the vacuum system. At MSU cleanroom, there is a RIE machine, Nordson March
RIE-1701(shown in Figure A.2).

Figure A.2 Nordson March RIE-1701.
Descum process steps are as follows:
1. Set parameters on a RIE front panel described below.
Table A.1 Set parameters of a RIE
Parameters

O2 flow rate

Power

Pressure

Time

Set values

10 sccm

120 W

30 mTorr

30 seconds

2. After setting parameters, press ‘Vacuum’ to pump down until the pressure goes to the
setting pressure.
3. Press ‘Gas’ button and turn to green light, and then press ‘RF’ to turn to green.
4. After 30 seconds for etching, then press ‘Purge’, and ‘Bleed’.
5. Press ‘Stop’ and ‘Chamber close’. And then, remove the sample from the chamber.
181

APPENDIX B
Indicator values of a W-band BWO to set the operating frequency
Table B.1 Indicator values of a BWO
Indicator value

Operating frequency

20.2

74.412 GHz

23.4

75.919 GHz

26.3

77.470 GHz

29.7

78.927 GHz

32.4

80.515 GHz

35

81.969 GHz

38.3

83.710 GHz

42.2

85.582 GHz

44.4

86.688 GHz

48

88.328 GHz

50.7

89.868 GHz

54.3

91.867 GHz

57.4

93.450 GHz

60.8

95.180 GHz

64.2

96.900 GHz

67.2

98.433 GHz

70.6

100.322 GHz

73.8

101.868 GHz

76.7

103.342 GHz

79.3

104.462 GHz

82.5

105.977 GHz

85.6

107.435 GHz

182

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