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thesis entitled

DESIGN AND ANALYSIS OF A WIDEBAND PATCH
ANTENNA FOR USE WITH A MINIATURE RADAR SYSTEM

presented by

NATHAN THOMAS KORNBAU

has been accepted towards fulfillment
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Master of degree in Electrical Engineering
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DESIGN AND ANALYSIS OF A WIDEBAND PATCH
ANTENNA FOR USE WITH A MINIATURE RADAR
SYSTEM

By

Nathan Thomas Kornbau

A THESIS

Submitted to
Michigan State University
in partial fulfillment of the requirements
for the degree of

MASTER OF SCIENCE
Department of Electrical Engineering

2008

ABSTRACT

DESIGN AND ANALYSIS OF A WIDEBAND PATCH ANTENNA
FOR USE WITH A MINIATURE RADAR SYSTEM
By

Nathan Thomas Kornbau

A radar system that operates in the high X-band and low-Ku band is being devel-
oped for use on a miniature unmanned air vehicle(Mini-UAV). For the system to meet
electrical and weight requirements a custom high bandwidth (15% or more), linearly
polarized, low profile antenna is required. Resonant microstrip patch antennas are
generally linearly polarized and have a low profile construction. However, like all res-
onant antennas, microstrip patch antennas have low bandwidth, usually around 3%
to 5%. Due to the other favorable characteristics of patch antennas, large amounts
of research have been performed to increase the their bandwidth. This thesis exam-
ines the use of an array of four identical passive patches symmetrically positioned
above a fifth driven patch antenna buried between two dielectric substrates. The
four patches are excited by coupling with the driven patch. The coupling between
the patches also results in an increase in bandwidth. Simulations using a commercial
finite element method solver were performed to find an antenna design to potentially
meet the Mini-UAV bandwidth requirements. The simulated design was then fabri-
cated and experimentally tested to verify the simulation results. The simulation and

experimental procedure and results will be presented in this thesis.

 

 

 

at

514’

 

ACKNOWLEDGMENTS

I would like to thank Dr. Leo Kempel, for the opportunity to work and study
at this institution. I would also like thank Dr. Ed Rothwell and Dr. Shanker Bala-
subramaniam for sharing their vast knowledge with me over the past couple of years.
I am grateful for the assistance and support provided by Dr. Jeff Berrie and Dave
Kuhl before and during the work performed for this research. Finally, thank you to

my parents, Tom and Millie, for their everlasting love and encouragement.

iii

TABLE OF CONTENTS

LIST OF TABLES ................................. vi
LIST OF FIGURES ................................ vii
CHAPTER 1
Introduction ..................................... 1
1.1 Motivation ................................. 1
1.2 Design Specifications ........................... 2
1.3 Overview .................................. 3
CHAPTER 2
Patch Antenna Theory ............................... 5
2.1 Microstrip Transmission Line Theory ..................
2.2 Patch Antenna Background Theory ................... 7
CHAPTER 3
Wideband Patch Antenna Design ......................... 18
3.1 Increasing Patch Antenna Bandwidth .................. 18
3.1.1 Effect of the Substrate ...................... 18
3.1.2 Effect of Coupled Feeds and Elements .............. 19
3.2 Broadbanding Method Selection ..................... 21
3.3 Material Selection ............................. 24
CHAPTER 4
Wideband Patch Antenna Simulations ...................... 26
4.1 HFSS Simulation Setup .......................... 26
4.1.1 Drawing the Antenna Geometry ................. 27
4.2 Simulating the Scaled Antenna ..................... 30
4.2.1 Port Impedance .......................... 30
4.2.2 Radiation Patterns ........................ 34
4.3 Antenna Tuning .............................. 36
4.3.1 Frequency Adjustment ...................... 37
4.3.2 Impedance Loop Size ....................... 38
4.3.3 Impedance Loop Position .................... 38
4.4 Tuned Antenna ............. t ................. 45
4.4.1 Simulated Impedance Results .................. 46
4.4.2 Simulated Radiation Patterns .................. 48

iv

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CHAPTER 5

Antenna Fabrication and Testing ......................... 55

5.1 Fabrication ................................ 55

5.1.1 Milling ............................... 55

5.1.2 Bonding .............................. 57

5.1.3 Attaching Feed .......................... 59

5.2 Testing ................................... 61

5.2.1 Single Port Impedance ...................... 61

5.2.2 Radiation Patterns ........................ 63
CHAPTER 6

Conclusions and Future Work ........................... 73

BIBLIOGRAPHY ................................. 76

LIST OF TABLES

Table 3.1 Original and scaled antenna dimensions. .............. 24

Table 4.1 Optimized antenna dimensions. ................... 45

vi

 

 

 

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EFFFFFFF

Figure 2.1
Figure 2.2
Figure 2.3

Figure 2.4.

Figure 2.5

Figure 2.6
Figure 2.7
Figure 2.8
Figure 3.1
Figure 3.2
Figure 4.1
Figure 4.2
Figure 4.3
Figure 4.4
Figure 4.5
Figure 4.6
Figure 4.7
Figure 4.8
Figure 4.9

Figure 4.10
Figure 4.11
Figure 4.12
Figure 4.13
Figure 4.14
Figure 4.15
Figure 4.16

Figure 4.17

LIST OF FIGURES

The geometry of a microstrip transmission line. .......... 6
The field sturcture of microstrip transmission line .......... 6

The geometry of a patch antenna fed by a microstrip transmission

line. .................................. 8
The side View of a patch antenna showing the electric field structure

underneath the patch. ........................ 9
The top View of a patch antenna showing the ”slots” created by

electric field fringing .......................... 10
View of xz-plane with the dots representing centers of two slots. . 15
E-plane patch antenna pattern from theory. ............ 17
H-plane patch antenna pattern from theory. ............ 17
The geometry of a patch fed by a coupled feed. .......... 20
The geometry of Legay’s stacked microstrip antenna. ....... 23
The antenna geometry drawn in HFSS ................ 28
Return loss of the simulated scaled antenna. ............ 31
VSWR of the simulated scaled antenna. .............. 32
Smith chart showing the impedance of the simulated scaled antenna. 34
Simulated scaled antenna E—Plane pattern at 12GHz. ....... 35
Simulated scaled antenna H-Plane pattern at 12GHz. ....... 36
VSWR plots of simulated antennas with varying L1. ....... 37
Smith Chart of a simulated antenna with y-spacing of 2mm. . . . 39
Smith Chart of a simulated antenna with y-spacing of 2.5mm. . . 39
Smith Chart of a simulated antenna with y-spacing of 3mm. . . . 40
Smith Chart of a simulated antenna with y-spacing of 3.5mm. . . 4O
VSWR of a simulated antenna with y-spacing of 2mm. ...... 41
VSWR of a simulated antenna with y—spacing of 2.5mm. ..... 41
VSWR of a simulated antenna with y-spacing of 3mm. ...... 42
VSWR of a simulated antenna with y-spacing of 3.5mm. ..... 42
Smith chart of a simulated antenna with 33/ = 3.67mm and W2 =

6.33. .................................. 43

Smith chart of a simulated antenna with Sy = 3mm and W2 = 5.67. 44

vii

FFFFF

Fi

 

 

Fi

Fi

Fl

 

Figure 4.18

Figure 4.19
Figure 4.20
Figure 4.21
Figure 4.22
Figure 4.23

Figure 4.24

Figure 4.25

Figure 4.26

Figure 4.27

Figure 4.28

Figure 4.29

Figure 4.30

Figure 4.31

Figure 4.32

Figure 4.33

Figure 4.34

Figure 5.1
Figure 5.2
Figure 5.3
Figure 5.4
Figure 5.5

Smith chart of a simulated antenna with Sy = 2.67mm and W2 =
5.33. .................................. 44

Smith chart of a simulated antenna with Sy = 2mm and W2 2 4.67. 45

Return loss of simulated optimized antenna design. ........ 46
Smith Chart of simulated optimized antenna design. ....... 47
VSWR of simulated optimized antenna design ............ 47
E—Plane Radiation pattern of simulated optimized antenna at.
llG'Hz. ................................ 48
E—Plane Radiation pattern of simulated optimized antenna at
11.5G’Hz. ............................... 49
E—Plane Radiation pattern of simulated optimized antenna at
12GHz. ................................ 49
E—Plane Radiation pattern of simulated optimized antenna at
12.5GHz. ............................... 50
E-Plane Radiation pattern of simulated optimized antenna at
13GHz. ................................ 50
E—Plane Radiation pattern of simulated optimized antenna at
13.5GHz. ............................... 51
H-Plane Radiation pattern of simulated optimized antenna at
llGH z. ................................ 52
H-Plane Radiation pattern of simulated optimized antenna at
11.5GH2. ............................... 52
H-Plane Radiation pattern of Simulated Optimized antenna at
12GHz. ................................ 53
H-Plane Radiation pattern of simulated optimized antenna at
12.5GHz. ............................... 53
H—Plane Radiation pattern of simulated optimized antenna at
13GHz. ................................ 54
H-Plane Radiation pattern of simulated optimized antenna at
13.5GHz. ............................... 54
Antenna being Milled. ........................ 56
Patch damaged by wrong size end mill ................ 58
Large areas of copper missed by mill ................. 59

Center conductor of SMA connector soldered to microstrip feedline. 60

Outer conductor of SMA connector soldered to antenna ground
plane.

Figure 5.6

Figure 5.7

Figure 5.8

Figure 5.9

Figure 5.10
Figure 5.11
Figure 5.12
Figure 5.13
Figure 5.14
Figure 5.15
Figure 5.16
Figure 5.17
Figure 5.18
Figure 5.19
Figure 5.20

Return loss of antenna from experimental data. .......... 62
VSWR of antenna from experimental data .............. 63
Smithchart from experimental data .................. 64
Antenna experimental E—plane gain at 12GHz. .......... 65
Antenna experimental E—plane gain at 13GHz. .......... 66
Antenna experimental E—plane gain at 14GHz. .......... 66
Antenna experimental E—plane pattern at 12GHz .......... 67
Antenna experimental E—plane pattern at 13GHz .......... 67
Antenna experimental E-plane pattern at 14GHz .......... 68
Antenna experimental H-plane gain at 12GH z. .......... 69
Antenna experimental H-plane gain at 13GH2. .......... 69
Antenna experimental H-plane gain at 14GHz. .......... 70
Antenna experimental H-plane pattern at 12GHz .......... 70
Antenna experimental H-plane pattern at 13GHz .......... 71
Antenna experimental H-plane pattern at 14GH z .......... 71

ix

 

 

 

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CHAPTER 1

INTRODUCTION

1 .1 Motivation

This research project was a collaborative effort between Michigan State University,
BerrieHill Research Corporation, Air Force Research Laboratories (AF RL) Sensors
Directorate to design an antenna for use with a miniature radar system that will
be mounted on an Air Force(USAF) Miniature Unmanned Air Vehicle(Mini-UAV).
Currently the Mini-UAV has an optical sensor that is capable of sending video to the
operator, however adding a small radar system will allow a larger field of view and
the ability of the Mini-UAV to detect objects through rain, fog, dust, and smoke all
of which hinder the usefulness of the optical sensor [4].

The miniature radar is currently in development at BerrieHill Research Corpora-
tion, the radar will have three modes of operation, Ground Moving Target Indica-
tion(GMTI), High range resolution(HRR), and Synthetic Aperture Radar (SAR). It
is being designed to operate in the high X and low Ku bands with a center frequency
of 12.5 GHz and a bandwidth of 1.9 GHz (15% bandwidth) with a Voltage Standing
Wave Ratio(VSWR) less than 2.

In addition to the radio frequency(RF) requirements there are also physical and
monetary restrictions. The Mini—UAV has a limited load carrying capacity so the
complete radar system must weigh less than 4139 and a total volume less than 20cm x

20cm x 10cm. It is also desired to keep the cost of the radar system low so in the case

that a Mini-UAV is lost or damaged there is a minimal monetary loss. The goal of
this thesis is to design, fabricate, and test an antenna that meets the RF and physical

requirements while keeping manufacturing and material costs to a minimum.

1.2 Design Specifications

As was presented in the previous section the antenna design must meet several criteria.
The RF or electrical requirements will be the most important requirements to meet.
In addition to the center frequency being at 12.5 GHz and a 15% bandwidth with a
VSWR less than 2, the antenna also needs to be linearly polarized. This will allow
the most energy possible to be re-absorbed by the antenna as the Mini—UAV moves
by keeping the antenna allgined withthe transmitted signal on its return trip. Also,
the pattern of the antenna should have only a single lobe.

The physical, manufacturing simplicity and monetary criteria are dependent on
one another. For example, a certain material will strongly influence the cost of the
antenna, the manufacturing ease, and the weight of the antenna. To meet all of these
requirements a resonant microstrip antenna design was selected. Microstrip antennas
are very simple mechanically, so can be manufactured using standard circuit board
processing techniques. Also, they are lightweight consisting only of sheets of dielectric
material and thin layers of copper. The dielectric material is relatively inexpensive
and easily obtainable from various manufacturers. Using a microstrip antenna design
also offers another advantage, the radar system requires a way of running signals from
the RF source to the antenna. The sources will be located behind the antenna so the

feeds will preferably be able to bend a certain degree. If a flexible dielectric material

 

 

 

n12

€l€

 

 

is this can easily be done by using microstrip transmission lines on the same piece of
material as the antenna, minimizing the use of connectors and cables.

The resonant microstrip antenna, or patch antenna, also meets several of the
electrical criteria as well. They can radiate lineraly polarized and the pattern of
patch antennas consists of a single lobe. However, a considerable draw back is that
patch antennas generally have a very limited bandwidth, around 2% to 5%, much less
than the bandwidth required by the radar system [3].

Due to the many positive features of patch antennas, low profile, size, and ease
of fabrication, much work has been done observing the methods of increasing their
bandwidth. Two methods are available to increase bandwidth, structural techniques
and a circuit theory approach [14]. In this thesis ideas from both of these methods
are used to design a resonant patch antenna that meets bandwidth. requirements of

the miniature radar system.

1.3 Overview

The second chapter will review the theory behind resonant patch antennas and basic
microstrip transmission lines. The third chapter will discuss methods of increasing
bandwidth of patch antennas and the general design used for this thesis.

Chapter 4 will introduce the simulation methods used for the antenna design.
The final part of Chapter 4 will show simulation results and identify trends that were
observed while the design was being tuned.

The process that was used to fabricate the antenna designs from Chapter 4 will

be shown in Chapter 5. This chapter will also describe the tests that were performed

on the fabricated antennas and present the results from the tests.
The final chapter will contain conclusions and recommendations for future work

based the research performed for this thesis.

CHAPTER 2

PATCH ANTENNA THEORY

Since this thesis examines the use of a variation of a basic patch antenna using
a microstrip transmission line as a feed, this chapter will first briefly review some
background theory behind basic microstrip transmission lines. Following this section

the theory behind a single rectangular microstrip patch antenna will be presented.

2.1 Microstrip Transmission Line Theory

Microstrip transmission lines will be the easiest way to feed a patch antenna, they
can be fabricated in the same way and at the same time as the patch antenna. The
basic geometry of of a microstrip transmission line is shown below in Figure 2.1. The
most basic way to analyze a microstrip transmission line is to assume the substrate
has a dielectric constant of one and use image theory with the ground plane to setup
a simple transverse electromagnetic(TEM) transmission line [9].

However, cases where the substrate’s dielectric constant is not one, especially
when there is air above the conducting material, are significantly more complicated.
In these cases a true TEM wave cannot exist due to the differing phase velocities in
each of the materials. If the substrate is electrically very thin, d << A, a quasi-TEM
wave exists where most of the field energy is located in the substrate with the highest
concentration below the conducting material as shown in Figure 2.2. In the quasi-
TEM case analysis can be used to provide accurate approximations of transmission

line impedance [9].

 

 

Figure 2.1. The geometry of a microstrip transmission line.

 

 

    
       

 
  

3“-

e:

.. S...tract

Figure 2.2. The field sturcture of microstrip transmission line.

The width of the conducting line, height of the substrate and the dielectric con-
stant of the substrate all contribute to the impedance of the transmission line. First,
an effective dielectric constant, 68, must be found since both air and the substrate

must be considered to find the line impedance(2.1).

cr+1+er—1 1
2 2 ,/1+12d/W

Using the effective dielectric in the following equations the characteristic

 

 

(2.1)

(e:

impedance of the transmission line can be found. There are two equations for cases

when the ratio of W/d is larger or smaller than one. For a ratio larger than one (2.2)

 

is used.
60 8d W
Z = 1 — — 2.2
0 M; 11 (W + 461) ( )
For a ratio smaller than one (2.3) is used.
120
Z, = W (2.3)

 

yam/d + 1.393 + 0.667ln(W/d + 1.444)]

A more detailed explanation of how there equations were obtained is presented in
[9]. These equations will be used later in the design stage of the antenna to find the

width needed to create a microstrip transmission line with an impedance of 509.

2.2 Patch Antenna Background Theory

A typical patch antenna consists of a dielectric material (substrate) sandwiched be-

tween a conducting patch and a conducting ground plane. Any shape of the conduct-

ing patch can be used, however, for this thesis a rectangular patch, similar to the
patch shown in Figure 2.3, is examined. The best way to analyze a patch antenna
is to treat the volume underneath the patch as a lossy resonant cavity where instead
of all the energy being dissipated in the dielectric material it is radiated from the
sides of the cavity. A patch antenna shares another properties with a resonant cavity,
typically both have a very narrow bandwidth of only a few percent. This narrow
bandwidth is a feature shared by all resonant type antennas. Also, like a cavity, the
dimensions of the patch determine the frequency at which the resonance occurs, the

frequency that radiates [13].

 

 

 

 

 

 

 

Gramd Plane

Figure 2.3. The geometry of a patch antenna fed by a microstrip transmission line.

As shown in Figure 2.3 this thesis will look at the case of a patch where L is

approximately half of a wavelength and is fed by a microstrip transmission line. The

Feed

       

      
 
  
    

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Figure 2.4. The side view of a patch antenna showing the electric field structure
underneath the patch.

half wavelength patch will allow the patch to resonant at the desired frequency and
have a real impedance. The feed will transport energy in to the volume underneath
the patch, here a standing wave will form as shown in Figure 2.4, similar to a parallel
plate capacitor. However, since the ends of the patch are open it can be seen that
some fringing occurs at the edges of the patch. These fringing fields cause the patch to
radiate. The fringing fields also extend the length of the patch electrically, this is why
L is approximately half of a wavelength. The following equation provides an estimate

of the physical value of L based on the wavelength, A, and dielectric constant, er, of

the substrate [13].

A

L = 0.49
ya

 

(2.4)

In the xz—plane the fields at the edges of the patch are identical in magnitude

however they have opposite z-components, thus cancel each other out.VV hen viewed

Slot 1 Slot 2

 

 

 

 

 

 

 

 

 

A, "’
«h, ""
E ., y Patch u.
.m' u.

., "b

Feed x w W

"b

q-D "D
"P "P
"P "P
"D "D
”D v)

v v

z
m
—f_
t—q
m

 

 

Figure 2.5. The top view of a patch antenna showing the ”slots” created by electric
field fringing.

from the top, Figure 2.5, it can be seen that in the xy-plane the electric fields have
the same phase, this causes radiation and the broadside pattern of patch antennas. In
cases where the substrate thickness is much thinner than a wavelength the width of
the fringing fields, 3, can be approximated to be the same as the substrate thickness.
The areas of the fringing fields can best be analyzed by being treated as slot apertures
of length 3 and and width 14". To show the field pattern of a patch antenna the pattern
from one of the two slots must be calculated and then array theory must be applied
to account for the second identical slot [7].

In order to solve the slot field patterns the electric fields in the slots, Ea, must
be converted to magnetic surface currents, M 3. This relationship is shown below in
(2.5) [5].

Ms=EaXfi (2.5)

Using the magnetic current the electric vector potential, F, can be found, (2.6).

E: CM/[SMS(TI)€$p(jfif-T’)dsl (2.6)

47rr

Substitute (2.5) into (2.6) results in the electric vector potential in terms of the electric

field in the area of the slot, Sa.

-. 313)
F=emn X//Sa Eaeatp(j/3r r")dS (2.7)

471'7‘

Horn Figure 2.5 it can be observed that Ea only has x-components, so Ea can be
written as E . = EOE. The magnitude of the electric field is represented by E0. Also,
in this situation ft in is the z-direction, 2. Making these substitutions in (2.7) results
in (2.8) [7].

E=cwzx X/fsa Eoieserflr r)dS S, (2.8)

47TT‘

Taking the cross product of (2.8).

1‘7.ng “exp—N (_j')8T //Sa e:(rpjl3r r)dS (2.9)

Based the the shape the of the slot the source coordinates should be chosen to be
in the cartesian system, 1" = :13'5: + y’g‘l. The radiation pattern is being solve for and
generally this is described in spherical coordinates, r” = i: sin 9 cos (I) + Qsin 9 sin <I> +
icose [5]. Applying these coordinate systems to the quantity in the exponential

yields 12 ' 'r' = 513’ sin 9 cos (I) + y' sin 6 sin (I). So the electric vector potential becomes

11

(am)

F = .E exp(—Jfir)

y 06 41m"

[[5 exp(jL3(a:IsinGcosT+yIsinOsinT))dSI (2.10)
(I

Now the limits of the integration can be entered into the equation and the integral
solved. The slot exists on the xy—plane so the surface integral will be over x and y.
The length of the slot, 3, is known to be approximately the thickness of the substrate.

The width of the slot is the same as the width of the patch antenna, W.

_. 14/2 2
F: QED €€—_—Ip47r_1]fir) /W/2./—://2 817p( (jL3( 1: IsinOcosT + y IsinOsin T))d:rI (13/

(2.11)

 

emp(—jL3r) Ws sin (é; sin 8 cos T) sin (Q3! sin 9 sin T)

F=gEe
0 47” @253- sin 6 cos T Q2”: sin 6 sin T

(2w)

Since 3 is much smaller than a wavelength and if the new variable u is set to equal

sinOcosT the first fraction term from the integral equation can fit the form of

(amnwl

lim sin—v = 1 (2.13)

v—>0 1)

Using (2.13) on (2.12), and converting 3) to spherical coordinates, (2.14), the
electric vector potential can be broken into ('9 and T components, (2.15) and (2.16)

respectively.

3}=fsinOsinT+OcosOsinT+TcosT (2.14)

12

sin (T sin 6 sin T)

 

, —- 3
Fe 2 cosOsin TEoch—Il/Vs

2.15
47”“ £2- sinesinT I I

exp(— —j[ir)W ssin (QTW sin 9 sin T)
47” QQIK sin 8 sin T

 

F4, = cos TEOC (2.16)

Now that the O and T components of the electric vector potential have been iden-
tified they can be used to find the magnetic field due to the electric vector potential,

but not the entire magnetic field [13].
If}? = —jw(Feé + Fqs‘I’) (2.17)

Since the radiation pattern is what is being solved for, the electromagnetic wave can
be approximated as a plane wave. In that case the electric field associated with
the electric vector potential can be related to the magnetic field associated with the

electric vector potential [5].
E} = anF x 1‘ = —jw77(Fq,T - Fee) (2.18)

Substitute (2.15) and (2.16) into (2.17)

earp(—jfir) Ws sin (Q2— sin 9 sin T)
47" EEK sin 9 sin T

 

E} = —jcw77E0 (cos TO — cosesin TT) (2.19)

If the ground plane that lies in the xy-plane is assumed to be perfectly conducting

13

image theory can be used to setup an equivalent but easier to solve problem. When
the problem is mirrored across the xy-plane, the electric currents reverse in direction,
and since the separation of the currents from the ground plane is very small, much
less than a wavelength, the electric currents cancel each other out. This eliminates
the need to solve for the magnetic vector potential. The magnetic currents do not
reverse direction, so the magnetic currents add together doubling their magnitude [5].

So total electric field from a slot can be shown as

:rp(—jL3r) WsSin (QZW— sin 9 sin T)

.. .. e A ..
E 2 2E = —j€w77E cos TO — cos O sin TT
T F 0 27" QZIZ sin 9 sin T I I

 

(2.20)
From (2.20) the principle plane patterns can be found by setting T to 00 for the

E—plane and T to 900 for the H-Plane. The normalized patterns are

GE—Plane(e) : 1 (2°21)

. W .
Sin @7— s1n O
GH—Plane(e) : C05 6 I ) (2'22)

'I—IQIZ sine

Now that the pattern for a single slot has been solved for, array theory must be

 

used to find the pattern from two slots. In order to find the array factor only the
center points of the two slots will be considered. For the case of the H-Plane Pattern
the points are always going to be equidistant so the energy received at any given

point will be doubled. Since the radiation patterns are normalized this will have no

14

effect on the H-plane pattern so

 

 

foe) = 1 (2.23)
{/20
{/2 086’
c089
\ 0
O G
[ L/2 —— lL/2 4

 

Figure 2.6. View of xz-plane with the dots representing centers of two slots.

The E—plane array factor is not as simple, as Figure 2.6 shows that any incoming
wave that has oblique incidence will arrive at the each of the two points with a
different phase. It can be seen that wave will arrive at the two points with a phase
difference of :l:L/ 2 sine depending on the side of the origin the point is on. From
this the array factor in the E—plane can be written as [7]

AFE = e$p(—jL3(L/2) sin 9) + e$p(jL3(L/2) sin 9) = 2 cos(,3% sin 9). (2.24)

15

After normalization the E—plane array factor is

fE = cos(fi—gi sin 9). (2.25)

Using the E and H-plane array factors with the total electric field found in (2.21)

and (2.22) the total fields of the array can be found.
L .
FE(9) : GE—Plane(@)fE(e) : (305035 5m 9) (2°26)

Sim/lg sin 9)
fig! sin 8

The fields found by (2.26) and (2.27) both have direct dependence on cosine of the

 

FH(9) = GH—Pzane(9)fH(@) = 0089 (227)

angle 9. So when the cosine term is maximized the pattern will be at a maximum.
This occurs at O = 0, along the z-axis. This is shown in Figures 2.7 and 2.8.

The theory behind patch antennas and microstrip has been presented in this
chapter. Chapter 3 will introduce some methods of increasing the bandwidth of
microstrip patch antennas and identify the an antenna type that will be further

investigated through simulations.

16

 

270

Figure 2.7. E—plane patch antenna pattern from theory.

 

270

Figure 2.8. H—plane patch antenna pattern from theory.

17

CHAPTER 3

WIDEBAND PATCH ANTENNA DESIGN

As was discussed in the last chapter microstrip patch antennas share several preper-
ties with resonant cavities, most importantly they are resonant for only a very limited
range of frequencies. This chapter will show several methods of increasing the band-
width of patch antennas and describe the approach that was used to design a patch

antenna meeting the bandwidth requirements for the Mini-UAV radar system.

3.1 Increasing Patch Antenna Bandwidth

Various way to increase the bandwidth of patch antennas have been researched. This
section will present a couple basic methods that are commonly used, more methods

can be found in [14].
3.1.1 Effect of the Substrate

The substrate used to build the patch antenna is a major contributor to the opera-
tional bandwidth of the antenna. Below is an approximation showing the relationship
of substrate dielectric constant, er, and thickness, t, to the bandwidth, BW, of the

antenna, assuming the thickness is much less than a wavelength, A [6].

II _ IE: (3.1)

BW=.
377 Er LA

 

Since the length, width, and wavelength of the antenna cannot be changed the only

tunable parameters of a rectangular patch antenna are the dielectric constant and

18

substrate thickness. The approximation suggests that a lower dielectric constant and
a thicker substrate will results in a wider bandwidth. In practical conditions using
a square microstrip antenna with a teflon substrate with a dielectric constant of 2.2
and a thickness where t / A = 0.02, the maximum bandwidth that can be expected is
about 2%, much less than what is required for the Mini-UAV radar.

The thickness of the substrate could be increased to improve bandwidth, but there
are undesirable effects when doing this. A thicker substrate allows higher order modes
to form under the patch degrading the radiation patterns, efficiency, and polariza-
tion characteristics. The thicker substrates also lead to increased radiation from the
microstrip feed line [14]. While the structural methods of increasing bandwidth are
important and will be implemented, other methods of increasing bandwidth without

having adverse effects on the antenna’s other characteristics are required.
3.1.2 Effect of Coupled Feeds and Elements

Another method for increasing the bandwidth of a patch antenna is to have the patch
fed through a coupled feed. Such a feed design will require two separate slabs of
substrate. As shown in Figure 3.1, the top of the upper slab will have the patch(solid
line), the feed line will sandwiched between the two slabs of substrate (broken line),
and the bottom of the lower slab will be the ground plane. The feed line will extend
underneath the patch by a distance S.

This feed configuration allows for several additional tunable parameters. There is
now an additional slab of substrate where the thickness and dielectric constant can

differ from the first slab. Also, the length that the feed line extends underneath the

19

 

 

 

_._.l,_.. W “a a]

 

 

 

 

 

Figure 3.1. The geometry of a patch fed by a coupled feed.

patch, S, can be tuned. The lower substrate can be selected for optimum microstrip
transmission line performance, thinner and higher dielectric constant. While the
upper slab can be selected for optimum antenna performance, thicker with a lower
dielectric constant. The length the feed line extends under the patch has an effect
on the reactance of the antenna input impedance, by varying the length of overlap
the impedance loop on the smith chart can be moved. For the best impedance match
over a certain bandwidth the largest number of frequency points as possible should
be as close as possible to the center of the smith chart [14].

Additional techniques for increasing bandwidth also exist. A stub can be placed
on the side of the feedline, the stub then acts and a capacitor in series with the
patch antenna. Adjustments to the side stub and the length that the feedline extends
underneath the patch can yield bandwidths as high as 20%. This increased bandwidth
comes at price of detriorated polarization characteristics and efficiency, since the side

stub acts as a radiating elements interfering with the patch’s radiation [14].

20

Coupling between antenna elements can also be used to increase the bandwidth
of the antenna. This approach looks at patches as coupled resonators which can be
tuned for Wideband operation using circuit theory. The impedance of the antenna can
be optimized by adjusting the spacing between elements, similar to the optimization
done to the length of the feedline discussed previously. Antenna elements can be
arranged in the same plane or stacked vertically.

The Mini-UAV requires a low profile antenna, so it would be desirable to use the
planar coupled patches. Unfortunately in this configuration the feed network quickly
becomes overly complex interfering with the antenna characteristics. Another draw
back of the planar patches is that the coupling is generally weak between patches
which results in narrower bandwidth. The patches can also be stacked vertically in
order to keep the feed network simple. The patches are separated by a slab of sub-
strate between each other. The thickness of this substrate provides another tunable
parameter. These stacked patches have a couple significant draw backs. The biggest
draw back of the vertical configuration is that the antenna is going to be taller than
a planar configuration, the taller antenna also requires more material increasing the

cost of the antenna [14].

3.2 Broadbanding Method Selection

The previous section showed methods of increasing bandwidth of patch antennas,
however no single way was capable of providing the required bandwidth while main-
taining a consistent pattern and polarization while meeting weight, ease of fabrication,

and cost requirements. In [8] a microstrip patch antenna is presented that maintains

21

a consistent pattern and polarization throughout a bandwidth as high as 25%. The
antenna is also light and relatively simple to fabricate requiring only two layers of
dielecric material and no vertical vias. This antenna design primarily employs the
coupled element method of increasing bandwdith, but substrate selection still plays
an important role in the antenna design.

Legay’s stacked microstrip antenna combines the methods of increasing bandwidth
described above into a single antenna design. The result is an antenna that has the low
profile of planar patches and the higher coupling and bandwidth of vertically stacked
patches. The general geometry of the antenna design is shown in Figure 3.2. The
lower level of conducting material(broken line) is between the two layers of dielectric
material. This level contained a microstrip line feeding a patch. The upper level
of conducting material(solid lines) contains four patches centered above the driven
patch on the lower level, these four patches are fed by coupling with the corners of
the driven patch. Due to the adjacent resonant modes excited on the coupled patches
and a broadbanding effect is realized [8].

It can be seen in Figure 3.2 that there are many parameters that can be tuned to
obtained the desired antenna characteristics. L1 and W1 are the lengths and widths
of the upper patches. L2 and W2 are the length and width of the lower patch. Sm and
83) are the distances between patches in both and x and y directions. The heights
of the two substrates can also be tuned for Optimal bandwidth. The width of the
microstrip line, W f , will depend on the thicknesses and dielectric constants of the
substrates.

This antenna design offers all the properties required for the Mini-UAV radar

22

 

L1

 

 

 

 

 

 

 

 

 

 

 

 

1" I
1_I——‘—————I
If: 5y W2
'_— 1

 

 

 

 

 

 

 

 

 

 

 

 

 

SX
[_ ..
X

Figure 3.2. The geometry of Legay’s stacked microstrip antenna.

system, bandwidths as high as 25%, linear polarization, single lobe radiation pattern,
low profile, light weight, low material cost, and relatively simple fabrication. The
antenna that was presented in [8] was designed to operate with a center frequency of
4.3 GHz, in order to use this type of antenna with the Mini-UAV radar system it must
be redeisgned to operate at 12.5 GHz. To do this the antenna design will be scaled
down in size. Table 3.1 shows the dimensions used in [8] and the resulting scaled
dimensions. The dimensions were scaled by determining the ratio of the wavelengths
of the two center freqeuncies (24mm/70mm z 34%) and reducing the dimensions
accordingly. These scaled values will be used only as approximate starting points,
they will need to be further tuned because a different substrate material will be used

for the Mini-UAV antenna.

23

 

 

 

 

 

 

 

 

 

Dimension Orginal Scaled
Frequency 4.3 GHz 12.5GHz
Wavelength 69.8mm 24mm
L1 20mm 6.88mm
W1 16mm 5.50mm
L2 21.3mm 7.32mm
W2 20mm 6.88mm
Sx 16mm 5.5mm
Sy 16mm 5.5mm

 

 

 

 

 

Table 3.1. Original and scaled antenna dimensions.

3.3 Material Selection

Before the antenna design process can proceed any further the materials that will be
used as substrates must be selected. In order to keep the construction simple the
same type and thickness of material will be used for both layers of substrate. One
of the requirements of the substrate material is that it should be flexible enough to
bend back about 900 to connect the transmission line feeds to the RF sources of the
radar system.

Originally Rogers 3003, a ceramic-filled polytetrafluoroethylene (PTFE, Teflon),
was selected because of it it its high flexibility and acceptable dielectric constant of
3 [11]. Unfortunately, the 3003 was not easily machine by an inexperienced operator.
In order to continue antenna design and testing an easier to machine material with
comparable electrical properties was selected, Rogers 4003. This material is a glass
reinforced hydrocarbon / ceramic laminate so it is not as flexible as the 3003, it also has
a higher dielectric constant of 3.5 [12]. The higher dielectric constant is not desirable

for a large bandwidth bandwidth antenna, but if an antenna can be designed to meet

24

the bandwidth requirements with a higher dielectric constant material it can be tuned
to operate with at least as large of a bandwidth using a material with a lower dielectric
constant. This is the material that will be used for the rest of the design, simulation,
fabrication, and testing stages. If it proves to be too rigid redesigning the antenna
with a slightly lower dielectric substrate will not be difficult.

A thickness of substrate material also needs to be selected. Rogers 4003 comes
in several standard thicknesses. As was shown earlier a thicker substrate has better
broadbanding effects. The thickest 4003 is readily available is 1.524mm (60mil),
however when two pieces of this thickness are stacked they the total thickness is
no longer much smaller than a wavelength, 2.4mm, so multiple modes will form
producing undesirable effects. The next step down has a thickness of 0.813mm (32
mil), when stacked this material is much smaller than a wavelength [12]. Also, Rogers
3003 is available in a thickness of 0.75mm (30 mil), so the thickness would not have
a large change if 3003 is required in the future [11].

Now that an antenna type that can meet the antenna requirements has been
identified, a design starting point has been made by sealing the original antenna
design, and a substrate has been selected the antenna tuning process can begin. The
antenna tuning will be performed primarily through simulations that will be discussed

in detail in Chapter 4.

25

CHAPTER 4

WIDEBAND PATCH ANTENNA SIMULATIONS

In the previous chapter, Chapter 3, a possible antenna design was identified, it was
also noted that due to changes in substrate material that simply scaling the antenna
design would not produce a good antenna design and additional tuning would be
required. This chapter will discuss how this tuning was accomplished using Ansoft
Corporation’s High Frequency Structural Simulator (HFSS). The tunable parameters
of the antenna will also be identified and their effects on the antenna’s electrical
properties will be shown. At the end of this chapter the dimensions of an antenna

capable of meeting the Mini-UAV requirements will be presented.

4.1 HFSS Simulation Setup

Ansoft’s HFSS uses the finite element method (FEM) to provide full wave electromag—
netic solutions to three dimensional geometries [2]. To use HFSS a three dimensional
model of the geometry that needs to be simulated must be drawn. Each three di-
mensional object in the model must also be assigned material properties such as
permittivity, loss tangent, and permeability. Once the model is drawn and the mate-
rial properties are assigned, HFSS creates a mesh of the model using basic geometric
shapes, usually triangles. FEM is then used to solve for the fields at the surfaces of
the modeled objects using the boundary conditions that are dependent on material
properties. The results from these calculations can then provide important informa-

tion such as port impedance, which can also be interpreted as VSWR, smith charts,

26

and return loss, and radiations patterns. Observations of both port impedance and

radiation patterns will be used to evaluate each iteration of the antenna design
4.1.1 Drawing the Antenna Geometry

The completed drawing of the antenna geometry can be seen in Figure 4.1. This
section will detail the steps taken to create the drawing. First a volume must be
defined in which the fields will be calculated. In this situation where, apart from the
antenna and its feed, everything is either freespace or ground plane, so a boundary box
can be constructed around where the antenna will be drawn. The bottom of the box
will be set to have an infinite ground plane boundary condition, the remaining 5 sides
will be configured as radiation boundaries acting as perfectly absorbing material. The
dimensions of the boundary box used for this simulation were chosen to be 26.67mm
x 26.67mm x 17.5mm, a volume large enough for the entire antenna to fit inside.
The bottom of the boundary box will be on the z = 0 plane, the top will be on the
z = 17.5mm plane. The box will also be centered at the origin of the xy-plane

Once the volume where the fields will be solved is defined, the various components
of the antenna can be drawn. Starting from the bottom the first component will be
a copper ground plane. The copper ground plane needs to cover the entire bottom
of the box, so it will be a 26.67mm square. A thickness of 35pm will used for the
copper, this is equivalent to a 102 / ft2 copper cladding that is commonly available
on dielectric materials. The copper square will be given the material properties of
copper in HFSS. The next component on the stack is the bottom slab for substrate

material, this also needs to entirely cover the ground plane so it is also a 26.67mm

27

 

Figure 4.1. The antenna geometry drawn in HFSS.

square. The substrate thickness was decided in Chapter 3 to be 0.81mm. The slab
is assign a dielectric constant of 3.5.

The transmission line feed and driven patch are placed on top of the bottom slab
of substrate. The L2 and W2 from Chapter 3 are used as the dimensions of the driven
patch. The thickness and material properties of the copper patch and transmission
line are the same as the ground plane, 35pm. The transmission line is chosen to
extend from the narrow edge of the driven patch on the negative x side to the edge
of the box on the negative x side. The width of the transmission line has not been
calculated yet.

The equations from Chapter 2, (2.1), (2.2), and (2.3), can be used as a. guideline,

but since the the transmission line is buried between two slabs of dielectric with air

28

on top of the upper slab the equations will not produce an accurate width. The
equations determine that the width of a uncovered 509 transmission line on 0.8lmm
Rogers 4003 should be 1.8mm. HF SS was then used to simulate a buried microstrip
transmission line with this width. The impedance was found to be too low, a couple
more simulations were run and a transmission line width of 1.6mm was found to have
have an impedance closest to 509, 49.29. This width was used in the simulation
along with the copper thickness of 35pm.

When building the antenna the two slabs of substrate will needed to be bonded,
this is accomplished through the use of a bonding film. More details about the
bonding film will be presented in Chapter 5. For the simulation only the thickness
and dielectric constant are required. The bonding film that will be used for bonding
the substrates is Rogers 3001, it has a thickness of 0.0381mm and a dielectric constant
of 2.28 [10]. The bonding film is also 26.67mm square.

The upper slab of substrate is the next component to be added to the stack, this
slab is identical in dimensions and properties to the bottom slab. The final layer
of the antenna stack contains the four top patches. The general layout of the four
patches in relation to the driven patch can be seen in Figure 3.2. The values for L1,
W1, SIL', and Sy are taken from Table 3.1. The thickness for this layer of copper is
the same as the other copper layer, 35pm.

The simulation is now almost ready to run, the last step is to setup an excitation
port. To do this a rectangle needs to be drawn on the side of the boundary box that
the transmission line touches, this will be the port where a signal would be connected

to the antenna. The port will be centered at y = 0 and have a width of five times

29

the width of the transmission line. In the z-direction port will extend from the top of
the ground plane to a height of five times the thickness of the lower slab of substrate.
The port must be this large to allow a realistic field structure to setup around the

transmission line.

4.2 Simulating the Scaled Antenna

While it has already been stated that due to changes in the substrate material the
scaled antenna would not have desirable performance, this section will present some
results of the simulation of the scaled antenna to Show that tuning does in fact need
to be performed. The simulation is run from 10GHz to 14GHz so the entire range

of frequencies the antenna needs operate is will be covered.
4.2.1 Port Impedance

The first plot, shown in Figure 4.2, displays the return loss of the antenna, or 811.
The return loss is a direct measurement of how much of the energy that is sent into
a device is returned to the source. In a short circuit all the energy is returned to
the source, a return loss of OdB. A perfectly matched matched source would have
no energy returned to the source. In reality this is not possible, but a return loss
of —40dB (1% of the energy returned) is considered good. This information can
be useful in determining what frequency the antenna is resonant/ radiates. A low
811 corresponds to less energy being reflected back to the source. In the case of an
antenna constructed out of low loss material the majority of the energy is going to
be radiated out into space at frequencies with a low return loss.

Figure 4.2 shows that there is a large and narrow dip in the return loss around

30

Scaled Patch Antenna 32mil 4003
S11

 

 

 

 

_5..

I
.3
0

Return Loss (dB)

 

 

 

 

-20 . "l
_25 1 1 1 i 1 1 1
10 10.5 11 11.5 12 12.5 13 13.5 14

Frequency (GHz)

Figure 4.2. Return loss of the simulated scaled antenna.

10.2GHz. This means that the antenna will radiate best at 10.2GH z, out of the range
required by the Mini-UAV. The narrow dip corresponds to a narrow bandwidth.
Often VSWR plots are used to show the bandwidth of an antenna. The lower
VSWR, the better the match between the antenna and source. A better match
results in more energy being radiated from the antenna and less energy be reflected
back to the source. Generally a VSWR of less than 2 is considered acceptable for

most applications, this corresponds to a return loss of about -9.54 dB or less. VSWR

can be calculated from 811 by (4.1).

1+10(—Sll/20)
VSWR = 1 _ 10(..511/20) (4-1)

 

31

The VSWR of the scaled antenna is shown in Figure 4.3. The vertical line in
Figure 4.3 represents a VSWR of 2. Frequencies where the VSWR is below this line
are where the antenna will be well matched with the antenna. Based on the range
of frequencies where the VSWR is below 2 the bandwidth of the antenna can be
calculated from (4.2). To find the bandwidth percentage, the center frequency of the
bandwidth must also be calculated, (4.3). Using bandwidth and center frequency the

bandwidth can be calculated, (4.4).

Scaled Patch Antenna 32mil 4003
VSWR
25 T I I I 1 l T

 

BandWidth: 0.5 GHZ (4.878%) Fc: 10.25 GHz
20 ~

15~

VSWR

10~

VSWR=2

 

 

 

 

l I

o l l l I L
10 10.5 11 11.5 12 12.5 13 13.5 14
Frequency (GHz)

 

Figure 4.3. VSWR of the simulated scaled antenna.

Bandwidth = H ighCrossingPoint — LowCrossingPoz'nt (4.2)

32

Bandwidth

 

CenterFrequency = 2 + LowCrossingPoz'nt (4.3)
Bandwidth
B d. 'dth = .
an wz % C ente'rF requency (4 4)

The results of these calculations are shown in Figure 4.3. The bandwidth is
500MHz (5%) centered at 10.25GHz. This only considers the frequencies from
IOGHz and up. Since 10.5GHz, the highest frequency with a VSWR less than 2,
is already still well below the frequency range of interest for this antenna, it can be
concluded that this design will not operate well in the required frequency range.

Another way to interpret the port impedance data is the smith chart. The smith
chart allows both the real and imaginary components of the input impedance of the
antenna to be plotted on a grid. The smith chart has many uses, however when
trying to match the antenna with the source impedance it can generally be stated
that as many frequency points as possible should be as close as possible to the center
of the chart. The center of the chart represents a perfect match to the reference
impedance, in this situation 509. The real axis crosses the center of the circle from
left to right. Points on the real axis to the left of center have a smaller real impedance
than reference, to the right, larger. The areas above and below the real axis represent
reactance, positive above and negative below. Figure 4.4 shows the smith chart data
from the scaled antenna.

From Figure 4.3 it was noted that the antenna had the best impedance match at
lower frequencies. This can be seen in Figure 4.4 as well, the lower frequency points

start out near the center of the smith chart. As frequency increases however the

33

Start:10 GHz
Stopz14 GHz

 

 

Figure 4.4. Smith chart showing the impedance of the simulated scaled antenna.

impedance of the antenna rapidly approaches the edges of the chart indicating a poor

match. This was also shown in the return loss and VSWR plots, Figures 4.2 and 4.3.
4.2.2 Radiation Patterns

From the port impedance provided by the simulation it is known that this scaled an—
tenna will not operate as desired in the frequency band of the Mini-UAV. Too much
of the signal fed to the antenna will be reflected back to the source, reducing the
amount that will be radiated. However, port impedance is not the only antenna pa-
rameter that needs to be considered, the radiation pattern of the antenna is also very
important. HFSS is capable calculating the far-field radiation pattern of geometries

it simulates.

34

Just because an antenna is radiating, does not mean that it is radiating equally
in all directions. Every antenna has a radiation pattern that will be determined by
how the electromagnetic fields behave in the geometry. In Chapter 2 the E-plane and
H-plane of a single patch antenna was derived, it indicated that the patch would have
a broadside pattern perpendicular to the plane of the patch. Since this antenna is
also a patch antenna it should also have a broadside pattern. Unfortunately, as can
be seen in Figures 4.5 and 4.6 the patterns of this antenna at 12GHz do not have
a main beam, the maximum gain observed in either of the two patterns is only a
fraction of a dB. As was stated previously the antenna is not well matched around

12GHz so not much energy is making it to the antenna to be radiated. From the

Scaled Patch Antenna 32mi| 4003
E-Plane, Max Gain: 0.44358d8

90 0.5

 

150‘

210'

270

Figure 4.5. Simulated scaled antenna E-Plane pattern at 12GHz.

35

Sealed Patch Antenna 32mil 4003
H-Plane. Max Gain: 0.43189dB

9° 0.5

 

150

210

270

Figure 4.6. Simulated scaled antenna H-Plane pattern at 12GHz.

results of the simulation of the scaled antenna it can be seen that this antenna will
not meet the performance requirements of the Mini-UAV radar system. This antenna
will be used as a starting point and the various parameters of the antenna will be

adjusted and the effects of these adjustments will be noted.

4.3 Antenna Tuning

In this section the various parameters of the antenna that can be tuned will be
presented and the results of tuning certain parameters will be discussed. The only
parameters that cannot be changes are the substrate height, substrate dielectric con-

stant, and the width of the transmission line feed. The sizes of all the patches and

36

the separation of the upper patches can all be adjusted. Many trials where performed

using HFSS, and trends were found that allowed for a design process to be developed.
4.3.1 Frequency Adjustment

The operating frequency of the simulated scaled antenna was noted as being lower
than desired. In Chapter 2 it was shown that the resonant frequency of a patch
antenna depends primarily on the lengths of the radiating patches. For this antenna
design the majority of the radiation comes from the top four patches. It can then be
reasoned that by changing the lengths of the top four patches, L1, that the frequency

the antenna operates at can be tuned. Figure 4.7 shows the VSWR of the simulated

 

 

 

 

 

   

6 .

--- L1 =5.5mm
1* L1=5.25mm ,/
5 " -—*— L1 =5mm I

I.

g 4 - /I -
a 3
3 ” if I

g‘

 

 

 

1 0 1 1 1 2 13 14
Frequency (GHz)

Figure 4.7. VSWR plots of simulated antennas with varying L1.

antennas with varying Ll’s. As expected a larger L1 results in a better match at

37

lower frequencies that have longer wavelengths. As L1 is decreased in size from

5.5mm to 5mm the best matched frequencies of the antenna increase from 10.7GHz

to 13.2GHz.
4.3.2 Impedance Loop Size

The impedance loop that can be observed on smith charts is a result from antenna
elements coupling together. As elements are closer together and they overlap the
driven patch more they have a stronger coupling, this results in larger coupling loop
on the smith chart. It was noticed that the y-spacing, Sy, had a much larger effect
than the x-spacing, Sr. This suggest a stronger coupling in the y-direction. Figures
4.8 to 4.11 show the effects of increasing the y-spacing from from 2mm up to 3.5mm.
As expected, the smaller spacings have larger impedance loop. It should also be noted
that changes in the y-spacing have almost not effect on the position of the loop.
The effect of the loop size on VSWR is shown in Figures 4.12 to 4.15. When the
impedance loop is larger (Figure 4.12) there are a greater number of frequency points
around the center, however none of the points are very close to the center. There
is no frequency where the VSWR is much less than 2. The smaller loops shown in
Figure 4.14 and 4.15 have a narrower range of frequencies near the center, but the

frequencies near the center are closer, resulting in a lower VSWR at those frequencies.
4.3.3 Impedance Loop Position

Apart from the changing the impedance loop size, the loop can be rotated. It has
already been shown that the area that the upper patches overlap the driven patch

controls the size of the impedance loop, if the overlap area is kept the same by

38

 

 

with y-spacing of 2mm.

Figure 4.8. Smith Chart of a simulated antenna

 

 

with y-spacing of 2.5mm.

Figure 4.9. Smith Chart of a simulated antenna

39

 

 

Figure 4.10. Smith Chart of a simulated antenna with y—spacing of 3mm.

 

 

Figure 4.11. Smith Chart of a simulated antenna with y-spaeing of 3.5mm.

40

4.5

 

3.5 -

VSWR
oo

 

2 \h /

 

 

110 111 12 1‘3 14

Frequency (GHz)

 

Figure 4.12. VSWR of a simulated antenna with y-spacing of 2mm.

 

4.5 . r

3.5

VSWR
o:

 

2
1.5. w

 

J

 

 

11 0 1 1 12
Frequency (GHz)

Figure 4.13. VSWR of a simulated antenna with y-spacing of 2.5mm.

41

13

14

4.5

 

3.5 ~

VSWR
o:

 

1.5'

 

l

 

 

11 0 1 1 1 2
Frequency (GHz)

Figure 4.14. VSWR of a simulated antenna with y-spacing of 3mm.

13

14

 

 

 

 

1.5-

 

L

 

 

11 0 1 1 1 2
Frequency (GHz)

Figure 4.15. VSWR of a simulated antenna with y—spacing of 3.5mm.

42

13

14

decreasing both the y-spacing and the width of the driven patch, W2, by the same
amount that the loop will rotate the smith chart counter-clockwise. By adjusting the
y-spacing and driven patch width the impedance loop can be moved to the center of
the smith chart, resulting in a better impedance match. Figures 4.16 through 4.19
show the counter clockwise movement of the smith chart as the y—spacing and driven

patch width is decreased.

 

Figure 4.16. Smith chart of a simulated antenna with S y = 3.67mm and W2 = 6.33.

43

 

5.67.

3mm and W2 =

Figure 4.17. Smith chart of a simulated antenna with 33/

 

= 5.33.

2.67mm and W2

Figure 4.18. Smith chart of a simulated antenna with Sy

44

 

110
120

130
140

70

N»

'u8 ‘°

\
. )-
i

V?!”

0.50

  
 
 

150

I

/

100 9°
1
o

y)

d

,99 1100 f as
an I

\l

-10

\\

I

21-»
7 ~30
- 4°
i”
o -00
"I10 —1oo .90 -oo '70

it

\

-100
—130
-12

Figure 4.19. Smith chart of a simulated antenna with Sy = 2mm and W2 = 4.67.

4.4 Tuned Antenna

Using the antenna tuning methods discussed above an antenna design that will poten-
tially meet the criteria for the Mini-UAV was found. Below are the simulation results

from the Optimized antenna design. The dimensions of this antenna are shown in

Table 4.1.

 

 

 

 

 

 

 

Dimension Optimized Value
L1 5.2mm
W1 4mm
L2 5.75mm
W2 4.67mm
Sx 4.25mm
Sy 3mm

 

 

 

 

Table 4.1. Optimized antenna dimensions.

45

4.4.1 Simulated Impedance Results

Figure 4.20 shows the simulated return loss of the antenna. It indicates the best
impedance match occurs around 12.6G'Hz. Also, acceptable match, where the return

loss is —10dB or less, ranges from 11GHz to 13.3GH2.

 

Return Loss (dB)

 

 

 

_20 1 1 at I i 1 1
10 10.5 11 11.5 12 12.5 13 13.5 14
Frequency (GHz)

 

Figure 4.20. Return loss of simulated Optimized antenna design.

The smith chart of the optimized design is shown in Figure 4.21. The impedance

loop is located near the center and has a midrange size, to reach the best compromise

of bandwidth and matching.

The VSWR of the design is presented in Figure 4.22. The simulation shows a

VSWR of less than two from 10.7GHz and 13.3GHz, 20.6% bandwidth.

46

Start210 GHz
Stop:14 GHz

 

 

Figure 4.21. Smith Chart Of simulated Optimized antenna design.

4.5

 

A
r

BandWidth: 2.5 GHz (20.5761 %) Fc: 12.15 GHz
3.5

VSVIR
u

N
at
Y

 

1.5_ \/

10 10.5 11 11.5 12 12.5 13 13.5 14
Frequency (GHz)

 

 

 

Figure 4.22. VSW’ R Of simulated Optimized antenna design.

47

4.4.2 Simulated Radiation Patterns

The simulated impedance results show the optimized antenna meets the bandwidth
requirements of the Mini-UAV. The radiation patterns of the antenna should also
be considered before the design is built and tested. Figures 4.23 to 4.28 show the
E—plane fields of the simulated optimized antenna design at frequencies from llGHz

to 13.5GHZ.

Optimized Antenna
E-Plane. Max Gain: 6.0479dB Q 11 GHz

9_° 8

 

270

Figure 4.23. E-Plane Radiation pattern of simulated Optimized antenna at llGHz.

In the E—plane the pattern of the antenna is a single broadside lobe at all frequen-

cies. The beam shifts a few degrees through the frequency sweep, but the shifts are

not large enough to be problematic. Since the transmission line feed is not symmetric

48

Optimized Antenna
E-Plane. Max Gain: 6.7149d8 0 11.5 Gl-lz

 

150
230

Figure 4.24. E—Plane Radiation pattern of simulated optimized antenna at 11.5GH z.

Optimized Antenna
E-Plane. Max Gain: 7.9646dB Q 12 GHz

90 8

270

Figure 4.25. E—Plane Radiation pattern of simulated optimized antenna at 12GHz.

49

Optimized Antenna
E-Plane, Max Gain: 8.5044dB G 12.5 GHz

9010

 

270

Figure 4.26. E—Plane Radiation pattern of simulated optimized antenna at 12.5GHz.

Optimized Antenna
E-Plane. Max Gain: 8.9806dB Q 13 GHz

9°10

 

270

Figure 4.27. E—Plane Radiation pattern of simulated optimized antenna at 13GHz.

50

Optimized Antenna
E-Plane. Max Gain: 9.3375dB Q 13.5 GHz

9010

 

 

270

Figure 4.28. E—Plane Radiation pattern of simulated Optimized antenna at 13.5GHz.

in the E-plane, it contributes to the radiation pattern and causes the pattern shifts.
The simulated gain of the antenna varies from 6dB to 9.3dB.

Figures 4.29 to 4.34 show the H-plane pattern through the same frequency sweep.
The contributions of the feedline are symmetric in the H-plane so main beam does
not shift in the H-plane.

The simulation shows that the optimized design has radiation pattern that offers
acceptable gain and shape through the entire bandwidth of the antenna. Based on the
simulation results of the Optimized antenna a prototype will be built. The fabrication

and testing of the prototype are presented in Chapter 5.

51

Optimized Antenna
H-Plane, Max Gain: 6.0479dB Q 11 GHz

 

1m ..

........
,,,,

180 .......... .......... ........ . _. 0

..,. .....

270
Figure 4.29. H-Plane Radiation pattern Of simulated Optimized antenna at llGHz.

Optimized Antenna
H-Plane. Max Gain: 6.7149d8 @ 11.5 GHz

9° 3

.....................

150 ..

., ..........

210 330

270

Figure 4.30. H-Plane Radiation pattern of simulated optimized antenna at 11.50Hz.

52

Optimized Antenna
I-I-Plane. Mm Gain: 7.6005d8 Q 12 GHz

 

150 ..

............
______

21o I ' I 330

270

Figure 4.31. H-Plane Radiation pattern Of simulated Optimized antenna at 12GHz.

Optimmd Antenna
H-Plane. Max Gain: 8.4761d8 O 12.5 GHz

9010

 

270

Figure 4.32. H-Plane Radiation pattern of simulated optimized antenna at 12.5GH2.

53

Optimized Antenna
H-Plane, Max Gain: 8.9386dB Q 13 GHz

9010

 

270

Figure 4.33. H-Plane Radiation pattern of simulated optimized antenna at 13GHz.

Optimized Antenna
H-Plane, Max Gain: 9.3375dB Q 13.5 GHz

9010

 

270

Figure 4.34. H-Plane Radiation pattern of simulated optimized antenna at 13.5GH2.

54

CHAPTER 5

ANTENNA FABRICATION AND TESTING

Chapter 4 used simulations to arrive at an antenna design that will, in simulation,
meet the requirements for the Mini-UAV radar system. To experimentally confirm
the simulation results prototypes will be fabricated. Impedance and radiation pattern
measurements will then we performed on the prototypes and the results compared
to simulation. This chapter will describe the fabrication and measurement processes

and present the experimental results.

5.1 Fabrication

The fabrication process starts with a sheet of dielectric material coated on each side
with a thin layer of copper. The copper will then be machined away leaving only
the patches and feedline. Since the antenna design requires two separate layers of
material the layers must also be bonded. After the machined material is bonded
together a standard RF connector will be connected to the transmission line so the

antenna can interface with RF test equipment.
5.1.1 Milling

The first step of the fabrication process is machining the dielectric material. Milling
is the standard machining method used to remove copper from a substrate. Other
methods do exist, however they require the use of hazardous chemicals and were not

used for this antenna.

55

A mill removes material to create geometries out of copper by moving a spinning
end mill across the plane of the material. The depth of the and mill is set to only
remove the copper and little to none of the substrate material. Modern mills are often
computer controlled by custom software interfacing that can import standard com-
puter aided drafting (CAD) files. For this antenna the antenna design was exported
from HFSS in the form of a. .dch file. The exported .dxf file was then modified using
AutoCAD, the area of the ground plane and substrate material was increased from
26.67mm square to 70mm x 78mm to allow at least a wavelength of substrate from
each patch to each edge. This was done so the antenna would behave more closely
to the simulated antenna with an infinite ground plane. Once the CAD files were
imported into the software, the material was placed on the milling table, and the end

mill depth set properly the milling was ready to start.

 

Figure 5.1. Antenna being Milled.

56

The process is automated requiring the user to only switch end mills and adjust
end mill depth if required. The end mills need to be switched in two situations.
First, as a mill is used the edges become dull and starts making rough cuts. Second,
different sized end mills are used for different situations. In areas of fine detail, around
patches and feedline, a smaller end mill should be used. If too large of an end mill is
used for detailed areas of the antenna can be damaged(Figure 5.2). For larger open
areas larger mills should be used to decrease the time spent machining the material.
For this antenna design a 10mil diameter end mill was used around the patches and
feedline and a 39mil diameter end mill was used for clearing out large open areas of
coppen

Once the copper surrounding the patches and feedlines has been removed the
remaining copper can also be peeled off using razors blades. This saves machine time
and can produce better results becasue The substrates do not always have uniform
thickness so large areas of copper are removed evenly by the end mill (Figure 5.3).
Once the two slabs of substrate have been successfully milled they are ready to be
bonded.

5.1.2 Bonding

For the bonding film to be properly applied it must be heated to 428°F and held at
that temperature for a minimum of 15 minutes while under 100psz' of pressure [10].
This is accomplished using a heated press. A piece of bonding film the same size as
the antenna is placed between slabs of substrate. The substrate slabs and bonding

film are aligned vertically placed in the press. In order to monitor the temperature a

57

 

Figure 5.2. Patch damaged by wrong size end mill.

58

 

Figure 5.3. Large areas of copper missed by mill.

thermocouple temperature sensor was positioned in the press near the antenna. The
press is then turned on and the heating process begins. At this time the antenna is
also put under 100ps'i of pressure. After the antenna is held at 428°F for 15 minutes
the heated is turned off and the antenna cools off under pressure. Finally, the cooled
off antenna can be removed from the press and is ready to have a. connector attached.

5.1.3 Attaching Feed

In order for measurements to be performed on the antenna a way to interface with
RF test equipment needs to be employed. For this antenna an SMA connector that is
designed to have a 500 impedance up to 18GHz was used [1]. The center conductor
of the SMA connector was soldered to the top of the microstrip feedline (Figure 5.4),

the outer conductor was soldered to the ground plane of the antenna(Figure 5.5).

59

 

Figure 5.4. Center conductor of SMA connector soldered to microstrip feedline.

 

Figure 5.5. Outer conductor of SMA connector soldered to antenna. ground plane.

60

5.2 Testing

Once the SMA connector is attached to the antenna a series of tests on the antenna
need to be performed to evaluate the antenna’s performance. First a single port
impedance test will be performed. This test will indicate how good of a match the
antenna has with the source. The impedance data can be plotted on smith charts
and VSWR plots, and bandwidth information can be found. Also, a series of pattern
tests will be performed using AFRL’S Radiation and Scattering Compact Antenna
Laboratory(RASCAL). The pattern test will indicate how well the antenna radiates

in different directions.
5.2.1 Single Port Impedance

AFRL’s Hewlett Packard 8510 Network Analyzer was used to measure the antenna’s
impedance. The analyzer was setup to measure the impedance of the antenna from
IOGHz to 15GH z. In order to calibrate the analyzer a single port short, open,
load method using a Hewlett Packard 8505B calibration kit was employed. During
the impedance measurements the antenna was positioned to radiate into a piece of
absorbing material to reduce its interaction with the environment. The results from
the impedance measurement are shown below.

Figure 5.6 shows the return loss of the antenna obtained from experimental testing.
It shows that the return loss of the antenna is less than -10dB from about 11.3GHz to
14GHz with the best match, —32dB, occurring at 11.7GHz. The simulated antenna
indicated a similar range of frequencies with a low return loss, 10.8C7Hz to 13.3GH z,

with a best match of —20dB occurring at 12.6GHz. There is some difference be-

61

4003 32m: Peeled Copper Film

 

Return Loss (dB)

 

 

 

10 11 12 13 14 15
Frequency (GHz)

Figure 5.6. Return loss of antenna from experimental data.

tween simulated and experimental results. The simulated antenna was assumed to
have an infinite ground plane, this obviously is not possible in practice. The simu-
lated antenna also did not include the SMA connector, the antenna could be having
some interactions with the connector. Fortunately, in this situation the experimental
data is slightly better for the Mini-UAV application. The center of the experimental
bandwidth, 12.65GHz, is closer to the design center frequency, 12.5G'Hz, than the
simulation, 12.25GH2. The experimental bandwidth is also higher, 21.5% compared
to the simulations 20.6%.

Figure 5.7 shows the VSWR of the tested antenna. VSWR is another way to
format the return loss data, so the same conclusions can be made from this data.

The best match, where the VSWR is closest to one, occurs at 11.60Hz and the

62

4003 32m" Peeled Copper Film
5.5 , ,

 

BandWidthz2.7063 GHz (21 .5048%)

3.5L, -

VSWR

2.. \\ x/ .

I

I
/
y/

 

l

 

 

l 4

13 14 15
Frequency (GHz)

 

1‘ 11
1.5 ~ \‘ ”/N\\~J”"N\\u/I
V
111 1'2

Figure 5.7. VSWR of antenna from experimental data.

antenna again is shown to have a bandwidth of 21.5%.

Figure 5.8 shows the smith chart of the measured impedance. The loops around
the center indicate the broadband match that was observed above. The smithchart
of the experimental data shows multiple loops around the center in contrast the sim-
ulated antennas single loop. This indicates the elements of the experimental antenna
coupling together slightly differently than in simulation, however it results in the

favorable characteristic of slightly increased bandwidth.
5.2.2 Radiation Patterns

The results from the impedance test show that the antenna will have the desired

bandwidth, but the radiation pattern of the antenna is also very important. AFRL’S

63

4003 32m" Pooled Copper Film
StarmO GHz
Stop:15 6112

all

I/

Vim.
‘ezs

’0
’6.
y/

'1.
J!
’fi

1

I

1

\
‘g‘lll
la

Figure 5.8. Smithchart from experimental data.

RASCAL was used to perform the radiation pattern measurements. RASCAL uses
an Agilent E83623 network analyzer and an Hewlett-Packard 8349B microwave am-
plifier to feed a linearly polarized horn antenna. The horn antenna can be rotated
to have either horizontal or vertical polarization. The waves from the horn antenna
are bounced off a reflector that is designed to convert the horn’s radiated waves into
plane waves directed at a target antenna. As the target antenna is rotated the plane
waves are used to find the far field pattern of the antenna. To test this antenna the
network analyzer was setup sweep from lOGH z to 15GHZ every 0.50 from 90° to
270°, where 180° corresponds to the 0° position of the antenna in the simulations.

The gain of the antenna was found by comparing its measured gain to the gain of a

calibrated horn antenna measured in RASCAL .

64

Since this antenna is linearly polarized antenna patterns of both horizontal(E—
plane) and vertical(H-Plane) polarization will be taken. The E-plane and H-plane
patterns of the antenna at 12GHz, 13GHz, and 14GHz will be shown in the following
figures. Figures 5.9 to 5.11 show the gain of E—plane of the antenna, the horizontal
line indicates —3dBi from maximum gain. Figures 5.12 to 5.14 Show the polar plots

of the E-plane pattern.

PATCH ARRAY at 1 ZGHz
32 mil 4003 FILM - PEELED Cu H-POL E-PLANE

j 4 I T

 

10 _ Max Gain:9.3026 dBi -1
5 ~ / A j./\/ " ‘ \\ .
//~ JV \
y
\ /f\\
f‘ 0 _ \ _// \\ -
(E i \
3 \
.E \
a -5 \ ‘
0 \

 

 

 

100 120 140 160 180 200 220 240 260
Position (Degrees)

Figure 5.9. Antenna experimental E—plane gain at 12GHz.

The experimental E—plane patterns show that there is not the simple lobe that the
simulations had predicted, Figures 4.23 to 4.28. The half power beam width of the

antenna in the E-plane is about 400. The peak gain of the patterns do all occur near

65

PATCH ARRAY at 13GHZ
32 mil 4003 FILM - PEELED Cu H.POL E-PLANE
15 I I I r I r I I I

 

Max Gain:11.1419 dBi

10r /i\\\ a
. I.“ . . . . \ .

Gain (dBI)
o

 

 

 

100 120 140 160 180 200 220 240 260
Position (Degrees)

Figure 5.10. Antenna experimental E—plane gain at 13GH z.

PATCH ARRAY at 14GHZ
32 mil 4003 FILM - PEELED Cu H-POL E-PLANE

I r I I I I I I I

 

10 Max Gain:9.4397 dBi .

/\/\\

g l
8 o;\ / \v”\q\ i
-5- / \ raj;
l M
U \/

 

 

100 120 140 160 180 200 220 240 260
Position (Degrees)

 

Figure 5.11. Antenna experimental E—plane gain at 14GHz.

66

PATCH ARRAY at 1ZGHz
32 mil 4003 FILM - PEELED Cu H-POL E-PLANE
Max Gain:9.3026dBi

 

 

270

Figure 5.12. Antenna experimental E—plane pattern at 12GHz.

PATCH ARRAY at 13GHZ
32 mil 4003 FILM - PEELED Cu H-POL E-PLANE
Max Gain:11.1419dBi

 

270

Figure 5.13. Antenna experimental E—plane pattern at 13GHz.

67

PATCH ARRAY at 14GHz
32 mil 4003 FILM - PEELED Cu H-POL E-PLANE
Max Gain:9.4397dBl

 

270

Figure 5.14. Antenna experimental E—plane pattern at 14GHz.

broadside so the antenna is radiating in the expected direction. The inconsistency
between simulation and experimentation could be partially explained by the use of
a finite ground plane and an SMA connector that were not simulated. Also, in
the E—plane of this antenna the feed line makes the antenna asymmetric about the
axis of rotation, so the feed line could be contributing to the radiation pattern. The
experimental gain of the antenna is around 2de' higher than simulation in this plane.
The feedline acting as a radiating element could account for this increase in gain.

Figures 5.15 to 5.17 show the gain of H-plane of the antenna, the horizontal line
indicates —3dBi from maximum gain. Figures 5.18 to 5.20 show the polar plots of
the H—plane pattern.

The experimental H-plane patterns show that there is a single lobe at broadside

68

 

PATCH ARRAY at 12GHz
32 mil 4003 FILM - PEELED Cu V-POL H-PLANE

j T TI T I I I I I

 

1O -
Max Gain:7.9194 dBi
fl.
//
5 » .. - // , \ .
/ \
/ \
//
0 F I] \ "
G
In
3 / \x
.S
5 / \ -
/ \\
/ t.
‘10 '- //\\/l \ 'i
/ x
-15 7 .

 

 

100 120 140 160 180 200 220 240 260
Position (Degrees)

 

Figure 5.15. Antenna experimental H—plane gain at 12GHz.

PATCH ARRAY at 13GHz
32 mil 4003 FILM - PEELED Cu V-POL H-PLANE

I I I I I l I I I

 

10 - Max Gain:8.9622 dBi 1

Gain (dBi)
/

 

 

 

100 120 140 160 180 200 220 240 260
Position (Degrees)

Figure 5.16. Antenna experimental H-plane gain at 13GH2.

69

 

PATCH ARRAY at 14GHz
32 mil 4003 FILM - PEELED Cu V-POL H-PLANE

I I I I I I I I I

10 ~ 1
Max Gain:7.2777 dBi

 

Gain (dBi)
\

40+ \ —

 

 

100 120 140 160 180 200 220 240 260
Position (Degrees)

 

Figure 5.17. Antenna experimental H-plane gain at 14GHz.

PATCH ARRAY at 1ZGHz
32 mil 4003 FILM - PEELED Cu V-POL H-PLANE
Max Gain:7.9194dBi

 

270

Figure 5.18. Antenna experimental H-plane pattern at 12GHz.

70

PATCH ARRAY at 13GHz
32 mil 4003 FILM - PEELED Cu V-POL H-PLANE
Max Gain:8.9622dBi

9° 1

 

270

Figure 5.19. Antenna experimental H-plane pattern at 13GHz.

PATCH ARRAY at 14GHz
32 mil 4003 FILM - PEELED Cu V-POL H~PLANE
Max Gain:7.2777dBi

9° 1

 

270

Figure 5.20. Antenna experimental H—plane pattern at 14GHz.

71

that the simulations had predicted, Figures 4.29 to 4.34. The half power beam width
of in the H-plane is around 80°. In the H-plane for this antenna the feedline is
along the axis of rotation, so it will always be contributing to the pattern equally.
The different orientations of the feedline and the resulting patterns indicate that the
feedline is probably radiating. The gain for the H-plane agrees within 0.3dBi for the
pattern in this plane.

This chapter has described the process of taking a simulated model on a computer
screen to an experimentally tested prototype. It was found that the simulated and
experimental results were often similar, however never exactly the same. Most of
the experimental results indicate that the antenna will Operate as required for the
Mini-UAV radar system. The experimental bandwidth of the antenna was found to
be over 21%, higher than the required 15%, and centered at 12.65GH2 not far from
the 12.5GHz radar center frequency. The H-plane of the antenna has a nice single
lobe pattern. However, the asymmetric E—plane pattern of the antenna does casue
some concern, some approaches that will be used to attmept to resolve this issue will

be presented in the following chapter.

72

CHAPTER 6

CONCLUSIONS AND FUTURE WORK

This thesis has produced a low profile(1.65mm), light weight, linearly polarized patch
antenna with 2.7GHz of bandwidth(21%) centered at 12.65GHz that meets the re-
quirements of the radar sytems being designed for AFRL’S Mini-UAV.

The physical requirements were meet by selecting a patch type of antenna that
easily lends itself to being low profile and light weight. The patch antenna is also
generally linearly polarized, but suffers from low bandwidth. In order to meet the
electrical requirements, previously discovered broadbanding techniques were used in
this thesis to arrive at a general high bandwidth antenna design. From the general de-
sign simulations were sued to tune the design. This tuned design was then fabricated
and experimentally tested.

The simulation and experimental results are similar in most cases, often times the
experimental results are slightly better. Simulations estimated a bandwidth around
20.6% centered at 12.250Hz, while experiments show 21.5% bandwidth centered at
12.65GHz. For the H—plane pattern, the simulated and experimental maximum gain
were within 1de' of each other, with a nice smooth, round lobe. The E—plane patterns
are where the simulated and experimental data start to differ substantially. The
experimental gain is around 2dBi more than the simulated. Also, the experimental
E—plane pattern had a uneven lobe that did not closely resemble the simulated data.

The shape of the patterns may have been affected by the test setup. Long cables

73

capable of introducing error were used and the antenna was not positioned in a test
body so interactions between the antenna and the mounting device were possible.

It is believed that deformed E-plane pattern is partially due a radiating feedline.
Future work will be performed in an attempt to smooth out the E—plane pattern by
reducing the radiation of the feedline. To do this a narrower feedline will be used. To
keep impedance of the feedline at 509 a thinner lower substrate material will also be
required. The thinner lower substrate will affect the dimensions of all the patches, so
the antenna will need to be further tuned using the new substrates.

Once a design with a smoother E—plane field is developed it will also be of interest
to transfer the design to a more flexible but more difficult to machine material, such
as Rogers 3003, to allow the feedline to be bent around to connect to the RF source
that will be located behind the antenna. Slight design changes will be required to
account for the lower dielectric constant of the new material.

The final improvement to the antenna design will be to position the final antenna
design into an array to improve gain. This will require additional tweaks to the patch

dimensions since there will be more antenna elements coupling together.

74

 

 

 

 

BIBLIOGRAPHY

75

BIBLIOGRAPHY

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[4] J. A. Berrie, Lightweight Miniature Sensor Payload for a Mini-UAV Phase I
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[5] R. F. Harrington, Time-Harmonic Electromagnetic Fields, New Jersey: John
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[6] D. R. Jackson and NC. Alexopoulos, ”Simple approximate Formulas for Input
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[10] Rogers Corporation, 3001 Bonding Film Properties and Laminating Techniques,
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[11] Rogers Corporation, R03000 Series High Frequency Circuit Materials, Data
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[12] Rogers Corporation, 1204000 Series High Frequency Circuit Materials, Data
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[13] W. L. Stutzman and G. A. Thiele, Antenna Theory and Design, Second ed. New
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76

Adi

 

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77 I

   

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